AD8337

General-Purpose, Low Cost, DC-Coupled VGA AD8337 FEATURES Low noise Voltage noise = 2.2 nV/√Hz Current noise = 4.8 pA/√Hz (positive input) Wide bandwidth (−3 dB) = 280 MHz Nominal gain range: 0 dB to 24 dB (preamp gain = 6 dB) Gain scaling: 19.7 dB/V DC-coupled Single-ended input and output High speed uncommitted op amp input Supplies: +5 V, ±2.5 V, or ±5 V Low power: 78 mW with ±2.5 V supplies FUNCTIONAL BLOCK DIAGRAM VPOS 8 AD8337 GAIN 7 GAIN CONTROL INTERFACE 18dB 1 VOUT PREAMP (PRA) INPP 3 + – 8 SECTIONS INPN 4 VCOM 2 5 6 PRAO VNEG APPLICATIONS Gain trim PET scanners High performance AGC systems I/Q signal processing Video Industrial and medical ultrasound Radar receivers Figure 1. GENERAL DESCRIPTION The AD8337 is a low noise, single-ended, linear-in-dB, generalpurpose variable gain amplifier (VGA) usable at frequencies from dc to 100 MHz; the −3 dB bandwidth is 280 MHz. Excellent bandwidth uniformity across the entire gain range and low output-referred noise makes the AD8337 ideal for gain trim applications and for driving high speed analog-todigital converters (ADCs). Excellent dc characteristics combined with high speed make the AD8337 particularly suited for industrial ultrasound, PET scanners, and video applications. Dual-supply operation enables gain control of negative-going pulses such as generated by photodiodes or photomultiplier tubes. The AD8337 uses the popular and versatile X-AMP® architecture, exclusively from Analog Devices, Inc., with a gain range of 24 dB. The gain control interface provides precise linear-in-dB scaling of 19.7 dB/V, referenced to VCOM. The AD8337 includes an uncommitted operational currentfeedback preamplifier (PrA) that operates in inverting or noninverting configurations. Using external resistors, the device can be configured for gains of 6 dB or greater. The AD8337 is characterized by a noninverting PrA gain of 2× using two external 100 Ω resistors. The attenuator has a range of 24 dB, and the output amplifier has a fixed gain of 8× (18.06 dB). The lowest nominal gain range is 0 dB to 24 dB and can be shifted up or down by adjusting the preamp gain. Multiple AD8337s can be connected in series for larger gain ranges, and for interstage filtering to suppress noise and distortion, and for nulling offset voltages. The operating temperature range is –40°C to +85°C, and it is available in an 8-lead, 3 mm × 3 mm LFCSP. Rev. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2005-2007 Analog Devices, Inc. All rights reserved. 05575-001 AD8337 TABLE OF CONTENTS Features .............................................................................................. 1 Applications....................................................................................... 1 Functional Block Diagram .............................................................. 1 General Description ......................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 5 ESD Caution.................................................................................. 5 Pin Configuration and Function Descriptions............................. 6 Typical Performance Characteristics ............................................. 7 Test Circuits..................................................................................... 14 Theory of Operation ...................................................................... 18 Overview...................................................................................... 18 Preamplifier................................................................................. 18 VGA.............................................................................................. 18 Gain Control ............................................................................... 18 Output Stage................................................................................ 19 Attenuator.................................................................................... 19 Single-Supply Operation and AC Coupling ........................... 19 Noise ............................................................................................ 19 Applications..................................................................................... 20 Preamplifier Connections ......................................................... 20 Driving Capacitive Loads.......................................................... 20 Gain Control Considerations ................................................... 21 Thermal Considerations............................................................ 22 PSI (Ψ) ......................................................................................... 22 Board Layout............................................................................... 22 Outline Dimensions ....................................................................... 24 Ordering Guide .......................................................................... 24 REVISION HISTORY 2/07—Rev. A to Rev. B Changes to Figure 30, Figure 31, and Figure 32 ......................... 11 Changes to Single-Supply Operation and AC Coupling Section ..................................................................... 19 Moved Noise Section to Page........................................................ 19 Changes to Ordering Guide .......................................................... 24 6/06—Rev. 0 to Rev. A Updated Format..................................................................Universal Changes to Table 3.............................................................................6 Changes to Figure 22, Figure 25, and Figure 26 ......................... 10 Changes to Figure 39 and Figure 40............................................. 13 Changes to Figure 74 and Figure 75............................................. 23 Updated Outline Dimensions....................................................... 25 Changes to Ordering Guide .......................................................... 25 9/05—Revision 0: Initial Version Rev. B | Page 2 of 24 AD8337 SPECIFICATIONS VS = ±2.5 V, TA = 25°C, PrA Gain = +2, VCOM = GND, f = 10 MHz, CL = 5 pF, RL = 500 Ω, including a 20 Ω snubbing resistor, unless otherwise specified. Table 1. Parameter GENERAL PARAMETERS –3 dB Small Signal Bandwidth –3 dB Large Signal Bandwidth Slew Rate Input Voltage Noise Input Current Noise Noise Figure Output-Referred Noise Output Impedance Output Signal Range Output Offset Voltage DYNAMIC PERFORMANCE Harmonic Distortion HD2 HD3 HD2 HD3 HD2 HD3 Input 1 dB Compression Point Two-Tone Intermodulation Distortion (IMD3) Conditions VOUT = 10 mV p-p VOUT = 1 V p-p VOUT = 2 V p-p VOUT = 1 V p-p f = 10 MHz f = 10 MHz VGAIN = 0.7 V, RS = 50 Ω, unterminated VGAIN = 0.7 V, RS = 50 Ω, shunt terminated with 50 Ω VGAIN = 0.7 V (Gain = 24 dB) VGAIN = −0.7 V (Gain = 0 dB) DC to 10 MHz RL ≥ 500 Ω, VS = ± 2.5 V, + 5 V RL ≥ 500 Ω, VS = ± 5 V VGAIN = 0.7 V (Gain = 24 dB) VGAIN = 0 V, VOUT = 1 V p-p f = 1 MHz f = 10 MHz f = 45 MHz VGAIN = −0.7 V, f = 10 MHz (preamp limited) VGAIN = +0.7 V, f = 10 MHz (VGA limited) VGAIN = 0 V, VOUT = 1 V p-p, f1 = 10 MHz, f2 = 11 MHz VGAIN = 0 V, VOUT = 1 V p-p, f1 = 45 MHz, f2 = 46 MHz VGAIN = 0 V, VOUT = 2 V p-p, f1 = 10 MHz, f2 = 11 MHz VGAIN = 0 V, VOUT = 2 V p-p, f1 = 45 MHz, f2 = 46 MHz VGAIN = 0 V, VOUT = 1 V p-p, f = 10 MHz VGAIN = 0 V, VOUT = 1 V p-p, f = 45 MHz VGAIN = 0 V, VOUT = 2 V p-p, f = 10 MHz VGAIN = 0 V, VOUT = 2 V p-p, f = 45 MHz VGAIN = 0.75 V, VIN = 50 mV p-p to 500 mV p-p 1 MHz < f < 100 MHz, full gain range Min Typ 280 100 625 490 2.15 4.8 8.5 14 34 21 1 VCOM ± 1.3 VCOM ± 3.8 ±5 Max Unit MHz MHz V/μs V/μs nV/√Hz pA/√Hz dB dB nV/√Hz nV/√Hz Ω V V mV −25 +25 Output Third-Order Intercept Overload Recovery Group Delay Variation −72 −66 −62 −63 −58 −56 8.2 −9.4 −71 −57 −58 −45 34 28 35 26 50 ±1 dBc dBc dBc dBc dBc dBc dBm dBm dBc dBc dBc dBc dBm dBm dBm dBm ns ns Rev. B | Page 3 of 24 AD8337 Parameter DYNAMIC PERFORMANCE Harmonic Distortion HD2 HD3 HD2 HD3 HD2 HD3 Input 1 dB Compression Point Two-Tone Intermodulation Distortion (IMD3) Conditions VS = ±5 V VGAIN = 0 V, VOUT = 1 V p-p f = 1 MHz f = 10 MHz f = 35 MHz VGAIN = −0.7 V, f = 10 MHz VGAIN = +0.7 V, f = 10 MHz VGAIN = 0 V, VOUT = 1 V p-p, f1 = 10 MHz, f2 = 11 MHz VGAIN = 0 V, VOUT = 1 V p-p, f1 = 45 MHz, f2 = 46 MHz VGAIN = 0 V, VOUT = 2 V p-p, f1 = 10 MHz, f2 = 11 MHz VGAIN = 0 V, VOUT = 2 V p-p, f1 = 45 MHz, f2 = 46 MHz VGAIN = 0 V, VOUT = 1 V p-p, f = 10 MHz VGAIN = 0 V, VOUT = 1 V p-p, f = 45 MHz VGAIN = 0 V, VOUT = 2 V p-p, f = 10 MHz VGAIN = 0 V, VOUT = 2 V p-p, f = 45 MHz VGAIN = 0.7 V, VIN = 0.1 V p-p to 1 V p-p −0.7 V < VGAIN < −0.6 V −0.6 V < VGAIN < −0.5 V −0.5 V < VGAIN < +0.5 V 0.5 V < VGAIN < 0.6 V 0.6 V < VGAIN < 0.7 V −0.6 V < VGAIN < +0.6 V VGAIN = 0 V No foldover −0.7 V < VGAIN < +0.7 V 24 dB gain change VPOS to VNEG (dual- or single-supply operation) Each supply (VPOS and VNEG) No signal, VPOS to VNEG = 5 V VGAIN = 0.7 V, f = 1 MHz Each supply (VPOS and VNEG) No signal, VPOS to VNEG = 10 V VGAIN = 0.7 V, f = 1 MHz 4.5 10.5 Min Typ Max Unit Output Third-Order Intercept Overload Recovery ACCURACY Absolute Gain Error −85 −75 −90 −80 −75 −76 14.5 −1.7 −74 −60 −64 −49 35 28 36 28 50 0.7 to 3.5 ±0.35 ±0.25 ±0.35 −0.7 to −3.5 19.7 24 12.65 −VS 70 0.3 200 5 15.5 78 −40 18.5 185 −40 10 23.5 +VS dBc dBc dBc dBc dBc dBc dBm dBm dBc dBc dBc dBc dBm dBm dBm dBm ns dB dB dB dB dB dB/V dB dB V MΩ μA ns V mA mW dB mA mW dB −1.25 −1.0 −1.25 +1.25 +1.0 +1.25 GAIN CONTROL INTERFACE Gain Scaling Factor Gain Range Intercept Input Voltage (VGAIN) Range Input Impedance Bias Current Response Time POWER SUPPLY Supply Voltage VS = ±2.5 V Quiescent Current Power Dissipation PSRR VS = ±5 V Quiescent Current Power Dissipation PSRR 13.5 25.5 Rev. B | Page 4 of 24 AD8337 ABSOLUTE MAXIMUM RATINGS Table 2. Parameter Voltage Supply Voltage (VPOS, VNEG) Input Voltage (INPx) GAIN Voltage Power Dissipation (Exposed Pad Soldered to PC Board) Temperature Operating Temperature Range Storage Temperature Range Lead Temperature (Soldering, 60 sec) Thermal Data—4-Layer JEDEC Board No Air Flow Exposed Pad Soldered to PC Board θJA θJB θJC ΨJT ΨJB Rating ±6 V VPOS, VNEG VPOS, VNEG 866 mW Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. –40°C to +85°C –65°C to +150°C 300°C ESD CAUTION 75.4°C/W 47.5°C/W 17.9°C/W 2.2°C/W 46.2°C/W Rev. B | Page 5 of 24 AD8337 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS VOUT VCOM INPP INPN 1 2 3 4 PIN 1 8 7 VPOS GAIN VNEG PRAO 05575-002 AD8337 TOP VIEW 6 (Not to Scale) 5 Figure 2. Pin Configuration Table 3. Pin Function Descriptions Pin No. 1 2 3 4 5 6 7 8 Mnemonic VOUT VCOM INPP INPN PRAO VNEG GAIN VPOS Description VGA Output. Common Ground when using Plus and Minus Supply Voltages. For single-supply operation, provide half the positive supply voltage at Pin VPOS to Pin VCOM. Positive Input to Preamplifier. Negative Input to Preamplifier. Preamplifier Output. Negative Supply (−VPOS for Dual-Supply; GND for Single-Supply). Gain Control Input Centered at VCOM. Positive Supply. Rev. B | Page 6 of 24 AD8337 TYPICAL PERFORMANCE CHARACTERISTICS VS = ±2.5 V, TA = 25°C, RL = 500 Ω, including a 20 Ω snubbing resistor, f = 10 MHz, CL = 2 pF, VIN = 10 mV p-p, noninverting configuration, unless otherwise noted. 30 25 20 15 10 5 0 –5 –800 % OF UNITS GAIN (dB) 60 +85°C +25°C –40°C 500 UNITS VGAIN = –0.4V 50 VGAIN = 0V VGAIN = +0.4V 40 30 20 10 05575-003 0 0 0.1 0.2 0.3 0.4 –0.5 –0.4 –0.3 –0.2 –0.1 0.5 –600 –400 –200 200 0 VGAIN (mV) 400 600 800 GAIN ERROR (dB) Figure 3. Gain vs. VGAIN at Three Temperatures See Figure 44 2.0 1.5 1.0 % OF UNITS Figure 6. Gain Error Histogram for Three Values of VGAIN 50 +85°C +25°C –40°C 40 500 UNITS –0.4V ≤ VGAIN ≤ +0.4V 0.5 GAIN (dB) 30 0 –0.5 –1.0 05575-004 20 10 05575-007 –1.5 –2.0 –800 0 19.3 19.4 19.5 19.6 19.7 19.8 19.9 GAIN SCALING (dB/V) 20.0 20.1 –600 –400 –200 0 200 VGAIN (mV) 400 600 800 Figure 4. Gain Error vs. VGAIN at Three Temperatures See Figure 44 2.0 1.5 1.0 0.5 GAIN (dB) Figure 7. Gain Scaling Histogram 50 RELATIVE TO BEST FIT LINE FOR 10MHz f = 1MHz f = 10MHz f = 70MHz f = 100MHz f = 150MHz 500 UNITS 40 0 % OF UNITS 30 –0.5 –1.0 05575-005 20 10 05575-008 –1.5 –2.0 –800 0 12.2 12.3 12.4 12.5 12.6 12.7 INTERCEPT (dB) 12.8 12.9 13.0 –600 –400 –200 200 0 VGAIN (mV) 400 600 800 Figure 5. Gain Error vs. VGAIN at Five Frequencies See Figure 44 Rev. B | Page 7 of 24 Figure 8. Intercept Histogram 05575-006 AD8337 30 25 20 GAIN (dB) 30 VGAIN = 0V VGAIN = +0.7 VGAIN = +0.5 VGAIN = +0.2 25 20 15 10 5 0 –5 100k GAIN (dB) VGAIN = 0 VGAIN = –0.2 15 10 5 0 –5 100k CL CL CL CL = 47pF = 22pF = 10pF = 0pF 1M 10M FREQUENCY (Hz) 100M VGAIN = –0.5 VGAIN = –0.7 1M 10M FREQUENCY (Hz) 100M 05575-009 500M 500M Figure 9. Frequency Response for Various Values of VGAIN See Figure 45 20 15 10 GAIN (dB) 10 Figure 12. Frequency Response for Three Values of CL with a 20 Ω Snubbing Resistor See Figure 45 VS = ±2.5V VS = ±5V VGAIN = +0.7 VGAIN = +0.5 VGAIN = +0.2 VGAIN = 0 8 5 0 VGAIN = –0.2 GAIN (dB) 6 VGAIN = –0.5 –5 VGAIN = –0.7 05575-010 4 2 05575-013 –10 –15 100k 1M 10M FREQUENCY (Hz) 100M 500M 0 100k 1M 10M FREQUENCY (Hz) 100M 500M Figure 10. Frequency Response for Various Values of VGAIN—Inverting Input See Figure 58 30 25 20 VGAIN = 0V 25 20 15 10 5 0 –5 –10 1M Figure 13. Frequency Response—Preamp See Figure 46 15 10 5 0 –5 100k CL = 47pF CL = 22pF CL = 10pF CL = 0pF 1M 10M FREQUENCY (Hz) 100M GROUP DELAY (ns) GAIN (dB) 05575-011 500M 10M FREQUENCY (Hz) 100M Figure 11. Frequency Response for Three Values of CL See Figure 45 Figure 14. Group Delay vs. Frequency See Figure 47 Rev. B | Page 8 of 24 05575-014 05575-012 AD8337 10 8 6 VS = ±5V 35 40 +85°C +25°C –40°C OFFSET VOLTAGE (mV) 4 NOISE (nV/√Hz) 05575-015 2 0 –2 –4 –6 –8 –10 –800 +85°C +25°C –40°C –600 –400 –200 0 VGAIN (mV) 200 400 600 VS = ±2.5V 30 25 20 05575-018 800 15 –800 –600 –400 –200 0 VGAIN (mV) 200 400 600 800 Figure 15. Offset Voltage vs. VGAIN at Three Temperatures See Figure 48 80 70 60 50 40 30 20 500 UNITS VGAIN = –0.4V VGAIN = 0V VGAIN = +0.4V Figure 18. Output-Referred Noise vs. VGAIN at Three Temperatures See Figure 50 25 +85°C +25°C –40°C 20 NOISE (nV/√Hz) % OF UNITS 15 10 5 05575-016 05575-019 10 0 –15 –10 –5 0 5 10 15 20 25 0 –800 –600 –400 –200 0 VGAIN (mV) 200 400 600 800 OUTPUT OFFSET VOLTAGE (mV) Figure 16. Output Offset Voltage Histogram for Three Values of VGAIN Figure 19. Short-Circuit, Input-Referred Noise at Three Temperatures See Figure 50 7 6 5 VGAIN = 0.7V RFB1 = RFB2 = 100Ω 1k VS = ±2.5V VS = ±5V 100 PREAMP GAIN = –1 IMPEDANCE (Ω) NOISE (nV/√Hz) 4 3 PREAMP GAIN = +2 2 1 0 100k 10 1 05575-017 0.1 1M 10M FREQUENCY (Hz) 100M 500M 1M 10M FREQUENCY (Hz) 100M Figure 17. VGA Output Impedance vs. Frequency See Figure 49 Figure 20. Short-Circuit, Input-Referred Noise vs. Frequency at Max Gain— Inverting and Noninverting Preamp Gain = −1 and +2 See Figure 50 Rev. B | Page 9 of 24 05575-020 AD8337 10 f = 10MHz, VGAIN = 0.7V –40 HD3 HD2 INPUT NOISE (nV/√Hz) INPUT REFERRED NOISE –50 DISTORTION (dBc) 1 –60 RS THERMAL NOISE ALONE –70 05575-021 0.1 1 10 100 1k –80 0 5 10 15 20 25 30 35 40 45 50 SOURCE RESISTANCE (Ω) LOAD CAPACITANCE (pF) Figure 21. Input-Referred Noise vs. RS See Figure 61 35 50Ω SOURCE 30 –40 –30 Figure 24. Harmonic Distortion vs. Load Capacitance See Figure 52 NOISE FIGURE (dB) 25 WITH 50Ω SHUNT TERMINATION AT INPUT UNTERMINATED 15 DISTORTION (dBc) –50 20 –60 10 05575-022 –70 5 –800 –600 –400 –200 0 VGAIN (mV) 200 400 600 800 –80 –800 –600 –400 –200 0 200 VGAIN (mV) 400 600 800 Figure 22. Noise Figure vs. VGAIN See Figure 51 –30 Figure 25. HD2 vs. VGAIN at Four Frequencies See Figure 52 –40 VOUT = 1V p-p VGAIN = 0V HD3 VS = ±2.5V HD3 VS = ±5V HD2 VS = ±2.5V HD2 VS = ±5V –40 1MHz 10MHz 35MHz 100MHz DISTORTION (dBc) DISTORTION (dBc) –50 –50 –60 –60 –70 05575-023 –70 05575-026 –80 0 200 400 600 800 1.0k 1.2k 1.4k LOAD RESISTANCE (Ω) 1.6k 1.8k 2.0k –80 –800 –600 –400 –200 0 200 VGAIN (mV) 400 600 800 Figure 23. Harmonic Distortion vs. RLOAD and Supply Voltage See Figure 52 Figure 26. HD3 vs. VGAIN at Four Frequencies See Figure 52 Rev. B | Page 10 of 24 05575-025 1MHz 10MHz 35MHz 100MHz 05575-024 AD8337 –30 VOUT = 2V p-p VOUT = 1V p-p VOUT = 0.5V p-p LIMITED BY MAXIMUM PREAMP OUTPUT SWING OUTPUT IP3 (dBm) 50 –40 40 DISTORTION (dBc) –50 30 –60 20 –70 –80 05575-027 10 VOUT = 1V p-p TONES SEPARATED BY 100kHz 0 –800 200 –600 –400 –200 0 VGAIN (mV) –90 –800 –600 –400 –200 0 200 VGAIN (mV) 400 600 800 400 600 800 Figure 27. HD2 vs. VGAIN for Three Levels of Output Voltage See Figure 52 –30 50 Figure 30. Output-Referred IP3 (OIP3) vs. VGAIN at Five Frequencies See Figure 64 –40 VOUT = 2V p-p VOUT = 1V p-p VOUT = 0.5V p-p LIMITED BY MAXIMUM PREAMP OUTPUT SWING OUTPUT IP3 (dBm) 40 DISTORTION (dBc) –50 30 –60 20 –70 –80 05575-028 10 VS = ±5V VOUT = 1V p-p TONES SEPARATED BY 100kHz –600 –400 –200 0 200 VGAIN (mV) 400 –90 –800 –600 –400 –200 200 0 VGAIN (mV) 400 600 800 0 –800 600 800 Figure 28. HD3 vs. VGAIN for Three Levels of Output Voltage See Figure 52 –20 Figure 31. Output-Referred IP3 (OIP3) vs. VGAIN, VS = ±5 V at Five Frequencies See Figure 64 20 15 10 IP1dB (dBm) –30 VOUT = 1V p-p VGAIN = 0V TONES SEPARATED BY 100kHz VS = ±2.5V VS = ±5V PREAMP LIMITED –40 IMD (dBc) 5 0 –5 –50 –60 –70 05575-029 –80 1M VS = ±2.5V VS = ±5V 10M FREQUENCY (Hz) 100M –15 –800 –600 –400 –200 200 0 VGAIN (mV) 400 600 800 Figure 29. IMD3 vs. Frequency See Figure 64 Figure 32. Input P1dB (IP1dB) vs. VGAIN See Figure 63 Rev. B | Page 11 of 24 05575-032 –10 05575-031 1MHz 10MHz 45MHz 70MHz 100MHz 05575-030 1MHz 10MHz 45MHz 70MHz 100MHz AD8337 80 60 40 20 VGAIN = 0.7V 8 6 4 2 800 600 400 200 CL = 0pF CL = 10pF CL = 22pF CL = 47pF 80 60 40 20 0 INPUT OUTPUT VS = ±2.5V VGAIN = 0.7V –10 0 10 20 30 TIME (ns) 40 50 60 VOUT (mV) VOUT (mV) VIN (mV) 0 –20 –40 –60 –80 –20 0 –2 –4 –6 0 –200 –400 –600 INPUT OUTPUT –20 –40 –60 05575-036 05575-037 –10 0 10 20 30 TIME (ns) 40 50 60 05575-033 –8 70 –800 –20 –80 70 Figure 33. Small Signal Pulse Response See Figure 53 80 60 INPUT 40 20 4 2 VGAIN = 0.7V 8 6 Figure 36. Large Signal Pulse Response for Three Capacitive Loads See Figure 53 800 600 400 200 CL = 0pF CL = 10pF CL = 22pF CL = 47pF 80 60 40 20 0 VOUT (mV) VOUT (mV) VIN (mV) 0 –20 –40 OUTPUT –60 –80 –20 0 –2 –4 –6 0 –200 –400 –600 INPUT OUTPUT VS = ±5V VGAIN = 0.7V –10 0 10 20 30 TIME (ns) 40 50 60 –20 –40 –60 –80 70 –10 0 10 20 30 TIME (ns) 40 50 60 05575-034 –8 70 –800 –20 Figure 34. Small Signal Pulse Response—Inverting Feedback See Figure 59 800 600 400 200 80 60 40 20 Figure 37. Large Signal Pulse Response for Three Capacitive Loads, VS = ±5 V See Figure 53 0.8 0.6 0.4 0.2 VOUT VGAIN = 0.7V VOUT (mV) (V) 0 –200 –400 –600 –800 –20 0 VIN (mV) 0 –0.2 –0.4 05575-038 INPUT OUTPUT –20 –40 –60 05575-035 –0.6 –0.8 –0.5 VGAIN 0 0.5 TIME (µs) 1.0 1.5 2.0 –10 0 10 20 30 TIME (ns) 40 50 60 –80 70 Figure 35. Large Signal Pulse Response See Figure 53 Figure 38. Gain Response See Figure 54 Rev. B | Page 12 of 24 VIN (mV) VIN (mV) AD8337 1.5 VGAIN = 0.7V VIN (V) VOUT (V) 10 0 –10 VGAIN = +0.7V, VS = ±2.5V VGAIN = +0.7V, VS = ±5V VGAIN = 0V, VS = ±2.5V VGAIN = 0V, VS = ±5V VGAIN = –0.7V, VS = ±2.5V VGAIN = –0.7V, VS = ±5V 1.0 0.5 PSRR (dB) –20 –30 –40 –50 –60 (V) 0 –0.5 –1.0 05575-039 05575-042 –70 –80 100k –1.5 –0.3 –0.1 0.1 0.3 0.5 0.7 TIME (µs) 0.9 1.1 1.3 1.5 1.7 1M 10M FREQUENCY (Hz) 100M Figure 39. Preamp Overdrive Recovery See Figure 55 1.5 VGAIN = 0.7V VIN (V) VOUT (V) QUIESCENT SUPPLY CURRENT (mA) Figure 42. PSRR vs. Frequency of Negative Supply See Figure 60 24 VS = ±5V VS = ±2.5V 1.0 22 0.5 20 (V) 0 18 –0.5 16 –1.0 05575-040 14 05575-043 –1.5 –0.3 –0.1 0.1 0.3 0.5 0.7 TIME (µs) 0.9 1.1 1.3 1.5 1.7 12 –50 –30 –10 10 30 50 TEMPERATURE (°C) 70 90 Figure 40. VGA Overdrive Recovery See Figure 56 10 0 –10 –20 PSRR (dB) Figure 43. Quiescent Supply Current vs. Temperature See Figure 57 VGAIN = +0.7V, VS = ±2.5V VGAIN = +0.7V, VS = ±5V VGAIN = 0V, VS = ±2.5V VGAIN = 0V, VS = ±5V VGAIN = –0.7V, VS = ±2.5V VGAIN = –0.7V, VS = ±5V –30 –40 –50 –60 –70 –80 100k 1M 10M FREQUENCY (Hz) 05575-041 100M Figure 41. PSRR vs. Frequency of Positive Supply See Figure 60 Rev. B | Page 13 of 24 AD8337 TEST CIRCUITS NETWORK ANALYZER NETWORK ANALYZER OUT 50Ω 50Ω IN OUT 50Ω 50Ω IN AD8337 3 AD8337 453 Ω 49.9Ω 4 + PRA – 20Ω 453Ω 1 3 49.9Ω 56.2Ω 5 7 4 + PRA – 1 20Ω 56.2Ω 5 7 100Ω VGAIN 100Ω 05575-044 100 Ω 100 Ω 05575-047 Figure 44. Gain and Gain Error vs. VGAIN Figure 47. Group Delay NETWORK ANALYZER FUNCTION GENERATOR OSCILLOSCOPE OUT 50Ω 50Ω IN OUT 50Ω CH1 50Ω 50Ω CH2 AD8337 3 453 Ω 3 4 7 VGAIN AD8337 + PRA – 1 DIFFERENTIAL FET PROBE 453 Ω 49.9Ω 4 + PRA – 20Ω 1 50Ω 5 7 100 Ω 100 Ω VGAIN 05575-045 Figure 45. Frequency Response Figure 48. Offset Voltage NETWORK ANALYZER NETWORK ANALYZER CONFIGURE TO MEASURE Z CONVERTED S22 OUT 50Ω 50Ω IN 50Ω IN 0Ω AD8337 3 NC 1 NC 3 AD8337 + PRA – 0Ω 1 49.9Ω 4 + PRA – 20Ω 453 Ω 49.9Ω 4 5 7 5 7 NC 100 Ω Figure 46. Frequency Response—Preamp Figure 49. Output Resistance vs. Frequency Rev. B | Page 14 of 24 05575-049 453Ω 05575-046 100 Ω 100Ω 100 Ω NC 05575-048 OPTIONAL POSITIONS FOR CLOAD 5 100 Ω 100 Ω AD8337 SPECTRUM ANALYZER PULSE GENERATOR OUT OSCILLOSCOPE POWER SPLITTER CH1 50Ω 50Ω CH2 IN 50Ω 0Ω AD8337 3 AD8337 3 49.9Ω 4 + PRA – 0Ω 1 4 + PRA – 1 20Ω 453Ω 49.9Ω 5 5 7 56.2Ω 7 100Ω 05575-050 100 Ω 100Ω VGAIN 0.7V 05575-053 100Ω Figure 50. Input-Referred and Output-Referred Noise Figure 53. Pulse Response DUAL FUNCTION GENERATOR OSCILLOSCOPE POWER SPLITTER NOISE FIGURE METER NOISE SOURCE DRIVE NOISE SOURCE 0Ω SINE WAVE SQUARE WAVE CH1 50Ω 50Ω CH2 INPUT 7 AD8337 49.9Ω (OR ∞) 3 4 AD8337 3 VGAIN DIFFERENTIAL FET PROBE + PRA – 1 0Ω 49.9Ω 4 + PRA – 20Ω 453 Ω 1 NC 5 7 5 100 Ω 05575-051 100Ω 05575-054 100 Ω VGAIN 100 Ω Figure 51. Noise Figure vs. VGAIN FUNCTION GENERATOR Figure 54. Gain Response OSCILLOSCOPE SPECTRUM ANALYZER SIGNAL GENERATOR INPUT 50Ω NC 7 RLOAD OUTPUT CH1 50Ω CH2 LOW PASS FILTER AD8337 3 AD8337 20Ω 1 3 49.9Ω 4 + PRA – 49.9Ω 4 + PRA – 1 NC CLOAD 5 5 7 100 Ω 100Ω 05575-055 100 Ω 100 Ω VGAIN 05575-052 100Ω Figure 52. Harmonic Distortion Figure 55. Preamp Overdrive Recovery Rev. B | Page 15 of 24 AD8337 FUNCTION GENERATOR POWER SPLITTER OUTPUT CH1 50Ω 50Ω CH2 OUT OSCILLOSCOPE OSCILLOSCOPE PULSE GENERATOR POWER SPLITTER CH1 50Ω 50Ω CH2 AD8337 3 49.9Ω 4 + PRA – 20Ω 453 Ω 1 AD8337 NC 3 4 5 100 Ω 100 Ω + PRA – 20Ω 453 Ω 1 56.2Ω 5 7 100Ω 05575-056 0.7V Figure 56. VGA Overdrive Recovery Figure 59. Pulse Response—Inverting Feedback +SUPPLY TO NETWORK ANALYZER BIAS PORT BENCH POWER SUPPLY DMM (+I) 8 NETWORK ANALYZER OUT 50Ω BYPASS CAPACITORS REMOVED FOR MEASUREMENT 1 IN 50Ω AD8337 3 4 VPOS + PRA – DMM (V) AD8337 3 49.9Ω 5 7 6 4 + PRA – 1 DIFFERENTIAL FET PROBE 100Ω 100 Ω DMM (–I) 05575-057 5 7 100 Ω 100 Ω 05575-060 VGAIN Figure 57. Supply Current Figure 60. PSRR NETWORK ANALYZER SPECTRUM ANALYZER OUT 50Ω 50Ω IN IN 50Ω AD8337 3 453Ω 3 4 AD8337 + PRA – 1 100Ω 100Ω 4 + PRA – 20Ω 1 5 7 5 7 VGAIN 05575-058 100Ω VGAIN Figure 58. Frequency Response—Inverting Feedback Figure 61. Input-Referred Noise vs. RS Rev. B | Page 16 of 24 05575-061 100Ω 100 Ω 05575-059 100Ω 100 Ω AD8337 SPECTRUM ANALYZER 22dB NETWORK ANALYZER POWER SWEEP IN 50Ω OUT 50Ω 50Ω IN 453Ω AD8337 3 4 + PRA – 3 AD8337 1 49.9Ω + PRA – 20Ω 1 4 5 100Ω 100Ω 7 0.7V 05575-062 5 7 100Ω 100 Ω VGAIN 05575-063 Figure 62. Short-Circuit Input Noise vs. Frequency Figure 63. IP1dB vs. VGAIN SPECTRUM ANALYZER INPUT 50Ω +22dB SIGNAL GENERATOR –6dB COMBINER –6dB 3 –6dB AD8337 453 Ω 49.9Ω + PRA – 20Ω 1 +22dB SIGNAL GENERATOR –6dB 4 5 7 100Ω 100 Ω 05575-064 VGAIN Figure 64. IMD and OIP3 Rev. B | Page 17 of 24 AD8337 THEORY OF OPERATION VPOS 8 RFB1 = RFB2 = 100Ω INPP RG INPN 4 3 + PRA 6dB – + ATTENUATOR –24dB TO 0dB – + 18dB (8X) – 749Ω 1 VOUT PRAO RFB2 RFB1 5 BIAS INTERPOLATOR GAIN INTERFACE 107 Ω VCOM 2 6 7 VNEG GAIN Figure 65. Block Diagram OVERVIEW The AD8337 is a low noise, single-ended, linear-in-dB, generalpurpose, variable gain amplifier (VGA) usable at frequencies up to 100 MHz. It is fabricated using a proprietary Analog Devices dielectrically isolated, complementary bipolar process. The bandwidth is dc to 280 MHz and features low dc offset voltage and an ideal nominal gain range of 0 dB to 24 dB. Requiring about 15.5 mA, the power consumption is only 78 mW from either a single +5 V or a dual ±2.5 V supply. Figure 65 is the circuit block diagram of the AD8337. VGA This X-AMP, with its linear-in-dB gain characteristic architecture, yields the optimum dynamic range for receiver applications. Referring to Figure 65, the signal path consists of a −24 dB variable attenuator followed by a fixed gain amplifier of 18 dB, for a total VGA gain range of −6 dB to +18 dB. With the preamplifier configured for a gain of 6 dB, the composite gain range is 0 dB to 24 dB. The VGA plus preamp with 6 dB of gain implements the following exact gain law dB ⎡ ⎤ Gain(dB) = ⎢19.7 ×V + ICPT (dB) GAIN ⎥ V ⎣ ⎦ PREAMPLIFIER An uncommitted, current-feedback op amp included in the AD8337 can be used as a preamplifier to buffer the ladder network attenuator of the X-AMP. As with any op amp, the gain is established using external resistors, and the preamplifier is specified with a noninverting gain of 6 dB (2×) and gain resistor values of 100 Ω. The preamplifier gain can be increased using larger values of RFB2, trading off bandwidth and offset voltage. The value of RFB2 should be ≥100 Ω because it and an internal compensation capacitor determines the 3 dB bandwidth, and smaller values can compromise preamplifier stability. Because the AD8337 is dc-coupled, larger preamp gains increase the offset voltage. The offset voltage can be compensated by connecting a resistor between the INPN input and the supply voltage. If the offset is negative, the resistor value connects to the negative supply. For ease of adjustment, a trimmer network can be used. For larger gains, the overall noise is reduced if a low value of RFB1 is selected. For values of RFB1 = 20 Ω and RFB2 = 301 Ω, the preamp gain is 16× (24.1 dB), and the input-referred noise is approximately 1.5 nV/√Hz. For this value of gain, the overall gain range increases by 18 dB; therefore, the gain range is 18 dB to 42 dB. where the nominal intercept (ICPT) is 12.65 dB. The ICPT increases as the gain of the preamp is increased. For example, if the gain of the preamp is increased by 6 dB, ICPT increases to 18.65 dB. Although the previous equation shows the exact gain law as based on statistical data, a quick estimation of signal levels can be made using the default slope of 20 dB/V for a particular gain setting. For example, the change in gain for a VGAIN change of 0.3 V is 6 dB using a slope of 20 dB/V and 5.91 dB using the exact slope of 19.6 dB/V. This is a difference of only 0.09 dB. GAIN CONTROL The gain control interface provides a high impedance input and is referenced to VCOM pin (in a single-supply application to midsupply at [VPOS + VNEG]/2 for optimum swing). When dual supplies are used, VCOM is connected to ground. The voltage on the VCOM pin determines the midpoint of the gain range. For a ground referenced design, the VGAIN range is from −0.7 V to +0.7 V with the most linear-in-dB section of the gain control between −0.6 V and +0.6 V. In the center 80% of the VGAIN range, the gain error is typically less than ±0.2 dB. The gain control voltage can be increased or decreased to the positive or negative rails without gain foldover. Rev. B | Page 18 of 24 05575-065 AD8337 The gain scaling factor (gain slope) is designed for 20 dB/V; this relatively low slope ensures that noise on the GAIN input is not unduly amplified. Because a VGA functions as a multiplier, it is important to make sure that the GAIN input does not inadvertently modulate the output signal with unwanted noise. Because of its high input impedance, a simple low-pass filter can be added to the GAIN input to filter unwanted noise. NOISE The total input-referred voltage and current noise of the positive input of the preamplifier are about 2.2 nV/√Hz and 4.8 pA/√Hz. The VGA output-referred noise is about 21 nV/√Hz at low gains. This result is divided by the VGA fixed gain amplifier gain of 8× and results in a voltage noise density of 2.6 nV/√Hz referred to the VGA input. This value includes the noise of the VGA gain setting resistors as well. If this voltage is again divided by the preamp gain of 2, the VGA noise referred all the way to the preamp input is about 1.3 nV/√Hz. From this, it is determined that the preamplifier, including the 100 Ω gain setting resistors, contributes about 1.8 nV/√Hz. The two 100 Ω resistors contribute 1.29 nV/√Hz each at the output of the preamp. With the gain resistor noise subtracted, the preamplifier noise is about 1.55 nV/√Hz. Equation 2 shows the calculation that determines the outputreferred noise at maximum gain (24 dB or 16×). where: • • • • • • • At is the total gain from preamp input to VGA output. RS is the source resistance. en − PrA is the input-referred voltage noise of the preamp. in − PrA is the current noise of the preamp at the INPP pin. en − en − RFB1 RFB 2 OUTPUT STAGE The output stage is a Class AB, voltage-feedback, complementary emitter-follower with a fixed gain of 18 dB, similar to the preamplifier in speed and bandwidth. Because of the ac-beta roll-off of the output devices and the inherent reduction in feedback beyond the −3 dB bandwidth, the impedance looking into the output pin of the preamp and output stages appears to be inductive (increasing impedance with increasing frequency). The high speed output amplifier used in the AD8337 can drive large currents, but its stability is susceptible to capacitive loading. A small series resistor mitigates the effects of capacitive loading (see the Applications section). ATTENUATOR The input resistance of the VGA attenuator is nominally 265 Ω. Assuming the default preamplifier feedback network RFB1 + RFB2 is 200 Ω, the effective preamplifier load is about 114 Ω. The attenuator is composed of eight 3.01 dB sections for a total attenuation range of −24.08 dB. Following the attenuator is a fixed gain amplifier with 8× (18.06 dB) gain. Because of this relatively low gain, the output offset is kept well below 20 mV over temperature; the offset is largest at maximum gain when the preamplifier offset is amplified. The VCOM pin defines the common-mode reference for the output, as shown in Figure 65. is the voltage noise of RFB1. is the voltage noise of RFB2. en − VGA is the input-referred voltage noise of the VGA (low gain, output-referred noise divided by a fixed gain of 8×). SINGLE-SUPPLY OPERATION AND AC COUPLING If the AD8337 is to be operated from a single 5 V supply, the bias supply for VCOM must be a very low impedance 2.5 V reference, especially if dc coupling is used. If the device is dc-coupled, the VCOM source must be able to handle the preamplifier and VGA dynamic load currents in addition to the bias currents. When ac coupling the preamplifier input, a bias network and bypass capacitor must be connected to the opposite polarity input pin. The bias generator for Pin VCOM must provide the dynamic current to the preamplifier feedback network and the VGA attenuator. For many single 5 V applications, a reference, such as the ADR391, and a good op amp provide an adequate VCOM source if a 2.5 V supply is unavailable. Assuming RS = 0 Ω, RFB1 = RFB2 = 100 Ω, At = 16×, and AVGA = 8×, the noise simplifies to en − out = (1.75 × 16)2 + 2(1.29 × 8)2 + (1.9 × 8)2 = 35 nV Hz (1) Dividing the result by 16 gives the total input-referred noise with a short-circuited input as 2.2 nV/√Hz. When the preamplifier is used in the inverting configuration with the same RFB1 and RFB2 = 100 Ω as previously noted, en − out does not change. However, because the gain dropped by 6 dB, the inputreferred noise increases by a factor of 2 to about 4.4 nV/√Hz. The reason for this increase is that the noise gain to the output of the noise generators stays the same, yet the preamp in the inverting configuration has a gain of −1 compared to the +2 in the noninverting configuration; this increases the input-referred noise by 2. R ) 2 + (en − RFB2 × A ) 2 + (e )2 × At ) 2 + (i × R ) 2 + (e × FB2 × A ×A en − out = (R × At ) 2 + (e n − RFB1 R VGA VGA n − VGA VGA S n − PrA n − PrA S FB1 (2) Rev. B | Page 19 of 24 AD8337 APPLICATIONS PREAMPLIFIER CONNECTIONS Noninverting Gain Configuration The AD8337 preamplifier is an uncommitted, current-feedback op amp that is stable for values of RFB2 ≥ 100 Ω. See Figure 66 for the noninverting feedback connections. INPP RG INPN PRAO GAIN (dB) DRIVING CAPACITIVE LOADS Because of the large bandwidth of the AD8337, stray capacitance at the output pin can induce peaking in the frequency response as the gain of the amplifier begins to roll off. Figure 68 shows peaking with two values of load capacitance using ±2.5 V supplies and VGAIN = 0 V. 25 VGAIN = 0V CL = 0pF CL = 10pF CL = 22pF 20 NO SNUBBING RESISTOR 15 PREAMPLIFIER 3 4 5 + – RFB2 RFB1 05575-066 10 Figure 66. AD8337 Preamplifier Configured for Noninverting Gain 5 Two surface-mount resistors establish the preamplifier gain. Equal values of 100 Ω configure the preamplifier for a 6 dB gain and the device for a default gain range of 0 dB to 24 dB. For preamp gains ≥2, select a value of RFB2 ≥ 100 Ω and RFB1 ≤ 100 Ω. Higher values of RFB2 reduce the bandwidth and increase the offset voltage, but smaller values compromise stability. If RFB1 ≤ 100 Ω, the gain increases and the inputreferred noise decreases. 0 05575-068 –5 100k 1M 10M FREQUENCY (Hz) 100M 500M Figure 68. Peaking in the Frequency Response for Two Values of Output Capacitance with ±2.5 V Supplies and No Snubbing Resistor 25 VGAIN = 0V CL = 0pF CL = 10pF 20 CL = 22pF WITH 20Ω SNUBBING RESISTOR 15 Inverting Gain Configuration For applications requiring polarity inversion of negative pulses, or for waveforms that require current sinking, the preamplifier can be configured as an inverting gain amplifier. When configured with bipolar supplies, the preamplifier amplifies positive or negative input voltages with no level shifting of the commonmode input voltage required. Figure 67 shows the AD8337 configured for inverting gain operation. Because the AD8337 is a very high frequency device, stability issues can occur unless the circuit board on which it is used is carefully laid out. The stability of the preamp is affected by parasitic capacitance around the INPN pin. Position the preamp gain resistors, RFB1 and RFB2, as close as possible to Pin 4, INPN, to minimize stray capacitance. INPP RFB1 INPN PRAO RFB2 05575-067 GAIN (dB) 10 5 0 05575-069 –5 100k 1M 10M FREQUENCY (Hz) 100M 500M Figure 69. Frequency Response for Two Values of Output Capacitance with a 20 Ω Snubbing Resistor PREAMPLIFIER 3 4 5 + – Figure 67. The AD8337 Preamplifier Configured for Inverting Gain Rev. B | Page 20 of 24 AD8337 In the time domain, stray capacitance at the output pin can induce overshoot on the edges of transient signals, as seen in Figure 70 and Figure 72. The amplitude of the overshoot is also a function of the slewing of the transient (not shown). The transition time of the input pulses used for Figure 70 and Figure 72 was set deliberately high at 300 ps to demonstrate the fast response time of the amplifier. Signals with longer transition times generate less overshoot. 800 600 400 200 80 60 40 20 0 800 VS = ±5V 600 400 200 60 40 20 0 INPUT OUTPUT CL = 0pF CL = 10pF CL = 22pF WITH 20Ω SNUBBING RESISTOR –10 0 10 20 30 40 TIME (ns) 50 60 70 80 –20 –40 –60 05575-073 80 VOUT (mV) 0 –200 –400 –600 –800 –20 –80 VOUT (mV) 0 CL = 0pF CL = 10pF CL = 22pF NO SNUBBING RESISTOR INPUT OUTPUT VIN (mV) Figure 73. Pulse Response for Two Values of Output Capacitance with ±5 V Supplies and a 20 Ω Snubbing Resistor –200 –20 –40 –60 05575-070 –400 –600 –800 –20 –10 0 10 20 30 40 TIME (ns) 50 60 70 –80 80 Figure 70. Pulse Response for Two Values of Output Capacitance with ±2.5 V Supplies and No Snubbing Resistor 800 600 400 200 80 60 40 20 0 The effects of stray output capacitance are mitigated with a small value snubbing resistor, RSNUB, placed in series with, and as near as possible to, the output pin. Figure 69, Figure 71, and Figure 73 show the improvement in dynamic performance with a 20 Ω snubbing resistor. RSNUB reduces the gain slightly by the ratio of RLOAD/(RSNUB + RLOAD), a very small loss when used with high impedance loads, such as ADCs. For other loads, alternate values of RSNUB can be determined empirically. The data for the curves in the Typical Performance Characteristics section of this data sheet are derived using a 20 Ω snubbing resistor. B B VOUT (mV) 0 VIN (mV) The best way to avoid the effects of stray capacitance is to exercise care in PC board layout. Locate the passive components or devices connected to the AD8337 output pins as close as possible to the package. Although a nonissue, the preamplifier output is also sensitive to load capacitance. However, the series connection of RFB1 and RFB2 is typically the only load connected to the preamplifier. If overshoot appears, it can be mitigated in the same way as the VGA output, by inserting a snubbing resistor. 05575-071 –200 INPUT OUTPUT CL = 0pF CL = 10pF CL = 22pF WITH 20Ω SNUBBING RESISTOR –10 0 10 20 30 40 TIME (ns) 50 60 70 –20 –40 –60 –80 80 –400 –600 –800 –20 GAIN CONTROL CONSIDERATIONS In typical applications, voltages applied to the GAIN input are dc or relatively low frequency signals. The high input impedance of the AD8337 enables several devices to be connected in parallel. This is useful for arrays of VGAs, such as those used for calibration adjustments. Under dc or slowly changing ramp conditions, the gain tracks the gain control voltage as shown in Figure 3. However, it is often necessary to consider other effects influenced by the VGAIN input. Figure 71. Pulse Response for Two Values of Output Capacitance with ±2.5 V Supplies and a 20 Ω Snubbing Resistor 800 VS = ±5V 600 400 200 60 40 20 0 INPUT OUTPUT CL = 0pF CL= 10pF CL = 22pF WITH NO SNUBBING RESISTOR –10 0 10 20 30 40 TIME (ns) 50 60 70 80 –20 –40 –60 05575-072 80 VOUT (mV) 0 –200 –400 –600 –800 –20 –80 Figure 72. Large Signal Pulse Response for Two Values of Output Capacitance with ±5 V Supplies and No Snubbing Resistor VIN (mV) Rev. B | Page 21 of 24 VIN (mV) AD8337 The offset voltage effect of the AD8337, as with all VGAs, can appear as a complex waveform when observed across the range of VGAIN voltage. Generated by multiple sources, each device has a unique VOS profile while the GAIN input is swept through its voltage range. The offset voltage profile seen in Figure 15 is a typical example. If the VGAIN input voltage is modulated, the output is the product of the VGAIN and the dc profile of the offset voltage, and it can be observed on a scope as a small ac signal as shown in Figure 74. In Figure 74, the signal applied to the VGAIN input is a 1 kHz ramp, and the output voltage signal is slightly less than 4 mV p-p. 10 8 6 VS = ±2.5V INPUT VS = 2 .5 OUTPUT Under certain circumstances, the product of VGAIN and the offset profile plus spikes is a coherent spurious signal within the signal band of interest and indistinguishable from desired signals. In general, the slower the ramp applied to the GAIN pin, the smaller the spikes are. In most applications, these effects are benign and not an issue. THERMAL CONSIDERATIONS The thermal performance of LFCSPs, such as the AD8337, departs significantly from that of leaded devices such as the larger TSSOP or QFSP. In larger packages, heat is conducted away from the die by the path provided by the bond wires and the device leads. In LFCSPs, the heat transfer mechanisms are surface-to-air radiation from the top and side surfaces of the package and conduction through the metal solder pad on the mounting surface of the device. θJC is the traditional thermal metric found in the data sheets of integrated circuits. Heat transfer away from the die is a 3dimensional dynamic, and the path is through the bond wires, leads, and the six surfaces of the package. Because of the small size of LFCSPs, the θJC is not measured conventionally; instead, it is calculated using thermodynamic rules. The θJC value of the AD8837 listed in Table 2 assumes that the tab is soldered to the board and that there are three additional ground layers beneath the device connected by at least four vias. For a device with an unsoldered pad, the θJC nearly doubles, becoming 138°C/W. OFFSET VOLTAGE (mV) 4 2 0 –2 –4 –6 –8 05575-075 –10 –800 –600 –400 –200 0 200 VGAIN (mV) 400 600 800 Figure 74. Offset Voltage vs. VGAIN for a 1 kHz Ramp The profile of the waveform shown in Figure 74 is consistent over a wide range of signals from dc to about 20 kHz. Above 20 kHz, secondary artifacts can be generated due to the effects of minor internal circuit tolerances, as seen in Figure 75. These artifacts are caused by settling and time constants of the interpolator circuit and appear at the output as the voltage spikes seen in Figure 75. 10 8 6 VS = ±2.5V INPUT VOUTPUT S = 2 .5 PSI (Ψ) Table 2 lists a subset of the classic theta specification, ΨJT (Psi junction to top). θJC is the metric of heat transfer from the die to the case, involving the six outside surfaces of the package. Ψ(XY) is a subset of the theta value and the thermal gradient from the junction (die) to each of the six surfaces. Ψ can be different for each of the surfaces, but since the top of the package is actually a fraction of a millimeter from the die, the surface temperature of the package is very close to the die temperature. The die temperature is calculated as the product of the power dissipation and ΨJT. Since the top surface temperature and power dissipation are easily measured, it follows that the die temperature is easily calculated. For example, for a dissipation of 180 mW and a ΨJT of 5.3°C/W, the die temperature is slightly less than 1°C higher than the surface temperature. OFFSET VOLTAGE (mV) 4 2 0 –2 –4 –6 –8 05575-074 SPIKE SPIKE BOARD LAYOUT –600 –400 –200 0 200 VGAIN (mV) 400 600 800 –10 –800 Figure 75. VOS Profile for a 50 kHz Ramp Because the AD8337 is a high frequency device, board layout is critical. It is very important to have a good ground plane connection to the VCOM pin. Coupling through the ground plane, from the output to the input, can cause peaking at higher frequencies. Rev. B | Page 22 of 24 AD8337 GND1 GND2 GND3 GND4 + C1 10µF L2 120nH 1 RVO3 0Ω J1 IN R4 0Ω R2 49.9Ω R5 RFB1 100Ω 2 3 4 8 7 6 5 CG 1nF R1 49.9Ω C3 0.1µF GAIN +VS –VS C2 10µF + L1 120nH VOUT RVO1 453Ω TP1 VOUT U1 VCOM INPP INPN RFB2 100Ω VPOS GAIN VNEG PRAO AD8337 C4 0.1µF RPO2 453Ω PRAO 05575-076 Figure 76. Evaluation Board Schematic—Noninverting Configuration 05575-077 Figure 77. Evaluation Board—Component Side Copper Figure 78. Evaluation Board—Wiring Side Copper Rev. B | Page 23 of 24 05575-078 AD8337 OUTLINE DIMENSIONS 3.00 BSC SQ 0.60 MAX 0.50 0.40 0.30 PIN 1 INDICATOR 8 1 PIN 1 INDICATOR TOP VIEW 2.75 BSC SQ 0.50 BSC (BOTTOM VIEW) EXPOSED PAD 1.50 REF 4 1.89 1.74 1.59 5 0.90 MAX 0.85 NOM 12° MAX 0.70 MAX 0.65 TYP 0.05 MAX 0.01 NOM 0.30 0.23 0.18 0.20 REF 1.60 1.45 1.30 EXPOSED PAD IS NOT CONNECTED INTERNALLY. FOR INCREASED RELIABILITY OF THE SOLDER JOINTS AND MAXIMUM THERMAL CAPABILITY IT IS RECOMMENDED THAT THE PAD BE SOLDERED TO THE GROUND PLANE. SEATING PLANE Figure 79. 8-Lead Lead Frame Chip Scale Package [LFCSP_VD] 3 mm × 3 mm Body, Very Thin, Dual Lead (CP-8-2) Dimensions shown in millimeters ORDERING GUIDE Model AD8337BCPZ-R2 1 AD8337BCPZ-REEL1 AD8337BCPZ-REEL71 AD8337BCPZ-WP1 AD8337-EVALZ1 AD8337-EVAL-INV AD8337-EVAL-SS 1 Temperature Range −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C Package Description 8-Lead Lead Frame Chip Scale Package [LFCSP_VD] 8-Lead Lead Frame Chip Scale Package [LFCSP_VD] 8-Lead Lead Frame Chip Scale Package [LFCSP_VD] 8-Lead Lead Frame Chip Scale Package [LFCSP_VD] Evaluation Board with Noninverting Gain Configuration Evaluation Board with Inverting Gain Configuration Evaluation Board with Single-Supply Operation Package Option CP-8-2 CP-8-2 CP-8-2 CP-8-2 Branding HVB HVB HVB HVB Z = Pb-free part. ©2005-2007 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05575-0-2/07(B) Rev. B | Page 24 of 24