ADL5502ACBZ-P7

450 MHz to 6000 MHz Crest Factor Detector ADL5502 FEATURES True rms response detector Envelope peak hold output Excellent temperature stability ±0.25 dB rms detection accuracy vs. temperature ±0.25 dB envelope detection accuracy vs. temperature; over the top 25 dB of the input range Over 35 dB input power dynamic range, inclusive of crest factor RF bandwidths from 450 MHz to 6 GHz Envelope bandwidths of 10 MHz 500 Ω input impedance Single-supply operation: 2.5 V to 3.3 V Low power: 3 mA at 3 V supply RoHS-compliant part FUNCTIONAL BLOCK DIAGRAM VPOS ADL5502 1kΩ 4nF INTERNAL FILTERING FLTR RMS CORE RFIN PEAK/ ENVELOPE ENBL BUFFERS 100Ω VRMS 100Ω PEAK CNTL COMM Figure 1. RF INPUT PEAK VRMS APPLICATIONS Power and envelope measurement of W-CDMA, CDMA2000, and QPSK-/QAM-based OFDM, and other complex modulation waveforms RF transmitter or receiver power and envelope measurement CNTL ENV TRACK 1µs/DIV PEAK HOLD 07631-002 Figure 2. GENERAL DESCRIPTION The ADL5502 is a mean-responding (true rms) power detector in combination with an envelope detector to accurately determine the crest factor of a modulated signal. It can be used in high frequency receiver and transmitter signal chains from 450 MHz to 6 GHz with envelope bandwidths over 10 MHz. Requiring only a single supply between 2.5 V and 3.3 V, the detector draws less than 3 mA. The input is internally ac-coupled and has a nominal input impedance of 500 Ω. The rms output is a linear-responding dc voltage with a conversion gain of 1.8 V/V rms at 900 MHz. The peak envelope output with a conversion gain of 1.2 V/V is toggled for peak hold with less than 1% output voltage droop in over 1 ms. The ADL5502 is a highly accurate, easy to use means of determining the rms and peak to the average value of complex waveforms. It can be used for crest factor measurements of both simple and complex waveforms but is particularly useful for measuring high crest factor (high peak-to-rms ratio) signals, such as W-CDMA, CDMA2000, and QPSK-/QAM-based OFDM waveforms. The peak hold function allows the capture of short peaks in the envelope with lower sampling rate ADCs. The crest factor detector operates from −40°C to +85°C and is available in an 8-ball, 1.5 mm × 1.5 mm wafer-level chip scale package. It is fabricated on a high fT silicon BiCMOS process. Rev. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2008–2011 Analog Devices, Inc. All rights reserved. 07631-001 ADL5502 TABLE OF CONTENTS Features .............................................................................................. 1  Applications ....................................................................................... 1  Functional Block Diagram .............................................................. 1  General Description ......................................................................... 1  Revision History ............................................................................... 2  Specifications..................................................................................... 3  Absolute Maximum Ratings............................................................ 6  ESD Caution .................................................................................. 6  Pin Configuration and Function Descriptions ............................. 7  Typical Performance Characteristics ............................................. 8  Circuit Description ......................................................................... 15  RMS Circuit Description and Filtering ................................... 15  Filtering ........................................................................................ 15  Envelope Peak-Hold Circuit ..................................................... 15  Output Buffers ............................................................................ 15  Measuring the Crest Factor ....................................................... 15  Applications Information .............................................................. 16  Basic Connections ...................................................................... 16  RF Input Interfacing................................................................... 16  Linearity....................................................................................... 17  Output Drive Capability and Buffering................................... 18  Selecting the Square-Domain Filter and Output Low-Pass Filter ............................................................................................. 18  Power Consumption, Enable, and Power-On/Power-Off Response Time ............................................................................ 19  Device Calibration and Error Calculation .............................. 19  Calibration for Improved Accuracy ......................................... 20  Calculation of Crest Factor (CF) .............................................. 20  Drift over a Reduced Temperature Range .............................. 21  Operation at High Frequencies ................................................ 22  Device Handling ......................................................................... 22  Evaluation Board ........................................................................ 23  Outline Dimensions ....................................................................... 25  Ordering Guide .......................................................................... 25  REVISION HISTORY 1/11—Rev. 0 to Rev. A Changes to Output Intercept Parameters, Table 1........................ 3 Changes to Figure 34 ...................................................................... 13 10/08—Revision 0: Initial Version Rev. A | Page 2 of 28 ADL5502 SPECIFICATIONS TA = 25°C, VS = 3.0 V, CFLTR = 10 nF, COUT = open, light condition ≤ 600 lux, 75 Ω input termination resistor, unless otherwise noted. Table 1. Parameter FREQUENCY RANGE RF INPUT (f = 900 MHz) Input Impedance RMS CONVERSION Dynamic Range ±0.25 dB Error 1 ±1 dB Error1 ±2 dB Error 2 Maximum Input Level Minimum Input Level Conversion Gain Output Intercept 3 Output Voltage, High Power In Output Voltage, Low Power In ENVELOPE CONVERSION Dynamic Range ±0.25 dB Error1 ±1 dB Error1 ±2 dB Error2 Maximum Input Level Minimum Input Level Conversion Gain Output Intercept3 Output Voltage, High Power In Output Voltage, Low Power In Test Conditions Input RFIN Input RFIN to output VRMS and PEAK No termination Input RFIN to output VRMS CW input, −40°C < TA < +85°C Min 450 Typ Max 6000 Unit MHz Ω||pF 330||1.04 ±0.25 dB error2 ±1 dB error2 VRMS = (Gain × VIN) + Intercept PIN = 5 dBm, 400 mV rms PIN = −15 dBm, 40 mV rms Input RFIN to output PEAK CW input, −40°C < TA < +85°C 27 33 29 12 −15 1.89 0.014 0.762 0.086 dB dB dB dBm dBm V/V rms V V V ±0.25 dB error2 ±1 dB error2 PEAK = (Gain × VIN) + Intercept PIN = 5 dBm, 400 mV rms PIN = −15 dBm, 40 mV rms 27 33 30 12 −15 1.27 0.014 0.516 0.062 dB dB dB dBm dBm V/V rms V V V Rev. A | Page 3 of 28 ADL5502 Parameter RF INPUT (f = 1900 MHz) Input Impedance RMS CONVERSION Dynamic Range ±0.25 dB Error 1 ±1 dB Error1 ±2 dB Error2 Maximum Input Level Minimum Input Level Conversion Gain Output Intercept3 Output Voltage, High Power In Output Voltage, Low Power In ENVELOPE CONVERSION Dynamic Range ±0.25 dB Error1 ±1 dB Error1 ±2 dB Error2 Maximum Input Level Minimum Input Level Conversion Gain Output Intercept3 Output Voltage, High Power In Output Voltage, Low Power In RF INPUT (f = 3500 MHz) Input Impedance RMS CONVERSION Dynamic Range ±1 dB Error1 ±2 dB Error2 Maximum Input Level Minimum Input Level Conversion Gain Output Intercept3 Output Voltage, High Power In Output Voltage, Low Power In ENVELOPE CONVERSION Dynamic Range ±1 dB Error1 ±2 dB Error2 Maximum Input Level Minimum Input Level Conversion Gain Output Intercept3 Output Voltage, High Power In Output Voltage, Low Power In Test Conditions Input RFIN to output VRMS and PEAK No termination Input RFIN to output VRMS CW input, −40°C < TA < +85°C Min Typ 238||0.90 Max Unit Ω||pF ±0.25 dB error2 ±1 dB error2 VRMS = (Gain × VIN) + Intercept PIN = 5 dBm, 400 mV rms PIN = −15 dBm, 40 mV rms Input RFIN to output PEAK CW input, −40°C < TA < +85°C 27 32 30 12 −15 1.75 0.010 0.700 0.079 dB dB dB dBm dBm V/V rms V V V ±0.25 dB error2 ±1 dB error2 PEAK = (Gain × VIN) + Intercept PIN = 5 dBm, 400 mV rms PIN = −15 dBm, 40 mV rms Input RFIN to output VRMS and PEAK No termination Input RFIN to output VRMS CW input, −40°C < TA < +85°C 26 32 30 12 −16 1.17 0.011 0.472 0.057 232||0.39 dB dB dB dBm dBm V/V rms V V V Ω||pF ±0.25 dB error2 ±1 dB error2 VRMS = (Gain × VIN) + Intercept PIN = 5 dBm, 400 mV rms PIN = −15 dBm, 40 mV rms Input RFIN to output PEAK CW input, −40°C < TA < +85°C 32 30 7 −16 1.52 0.002 0.594 0.065 dB dB dBm dBm V/V rms V V V ±0.25 dB error2 ±1 dB error2 PEAK = (Gain × VIN) + Intercept PIN = 5 dBm, 400 mV rms PIN = −15 dBm, 40 mV rms 32 31 7 −16 1.02 0.005 0.403 0.049 dB dB dBm dBm V/V rms V V V Rev. A | Page 4 of 28 ADL5502 Parameter VRMS OUTPUT Maximum Output Voltage Output Offset Pulse Response Time Available Output Current PEAK OUTPUT Maximum Output Voltage Output Offset Available Output Current Envelope Modulation Bandwidth Peak Hold Time CONTROL INTERFACE Logic Level to Track Envelope, High Input Current when High Logic Level for Peak Hold Condition, Low Enable Time ENABLE INTERFACE Logic Level to Enable Power, High Condition Input Current when High Logic Level to Disable Power, Low Condition Power-Up Response Time 4 POWER SUPPLIES Operating Range Quiescent Current Disable Current 6 1 2 3 Test Conditions Pin VRMS VS = 3.0 V, RLOAD ≥ 10 kΩ No signal at RFIN 10 dB step, 10% to 90% of settling level, no filter capacitor Pin PEAK VS = 3.0 V, RLOAD ≥ 10 kΩ No signal at RFIN Min Typ 2.4 15 15 3 1.5 14 3 10 600 Max Unit V mV μs mA V mV mA MHz μs V μA V μs V μA V μs μs V mA μA 100 100 5 1% voltage drop from last peak, CNTL = high 2.5 V ≤ VS ≤ 3.3 V, −40°C < TA < +85°C 2.5 V at CNTL, –40°C ≤ TA ≤ +85°C 2.5 V ≤ VS ≤ 3.3 V, −40°C < TA < +85°C 0 dBm at RFIN, CNTL held high for >1 μs Pin ENBL 2.5 V ≤ VS ≤ 3.3 V, −40°C < TA < +85°C 2.5 V at ENBL, –40°C ≤ TA ≤ +85°C 2.5 V ≤ VS ≤ 3.3 V, −40°C < TA < +85°C CFLTR = open, 0 dBm at RFIN CFLTR = 10 nF, 0 dBm at RFIN −40°C < TA < +85°C No signal at RFIN, 5 ENBL high input condition ENBL input low condition, CNTL in high condition 1.8 0.05 −0.5 1 ms without practically any drop in voltage. This circuit has the option of either transferring the envelope in real-time or in the peak-hold mode by toggling a control logic pin (CNTL). In the peak-hold mode, the output only is updated when a peak bigger than the previous biggest peak occurs. The PEAK output can be expressed as 2 PEAK T 1 = B × max[envelope (VIN )]T 1 T T2 RMS CIRCUIT DESCRIPTION AND FILTERING The rms processing is done using a proprietary translinear technique. This method is a mathematically accurate rms computing approach and allows achieving unprecedented rms accuracies for complex modulation signals irrespective of the crest factor of the input signal. An integrating filter capacitor does the square-domain averaging. The VRMS output can be expressed as T2 where: T1 is the time at which CNTL goes from high to low which is followed by a time where CNTL stays low. T2 is the time at which the PEAK measurement is taken, while CNTL is still low. Here again the only scaling parameter involved is a Scalar B, which is also decided on by the on-chip resistor ratio. OUTPUT BUFFERS A dual buffer takes in internal rms and envelope/peak signals and gains these up accordingly before these are brought out on the VRMS and PEAK pins. The output stage of the rms buffer is a common source PMOS with a resistive load to provide a railto-rail output. However, output stage of the PEAK buffer is an emitter-follower NPN stage with a resistive load to provide high speed characteristics for this output. This however limits the maximum voltage on the PEAK output to about 1.2 V below supply, resulting in a lower scaling factor for the PEAK signal path. Such a stage allows fast tracking of a rising transition when a very narrow peak is to be followed in the 10 MHz signal bandwidth. It is highly recommended that capacitive loads greater than 2 pF are avoided on the PEAK output to realize the full bandwidth potential of the device. Both the buffers have 100 Ω on-chip series resistances on the output. This allows for easy low-pass filtering of the two outputs. VRMS = A × ∫ V IN × dt 2 T1 T2 − T1 Note that A is a scaling parameter that is decided on by the on-chip resistor ratio, and there are no other scaling parameters involved in this computation, which means that the rms output is inherently free from any sources of error due to temperature, supply, and process variation. FILTERING The on-chip rms filtering is sufficient for most common handset applications, but an external filter capacitor can be connected if additional filtering is required; however, this increases the averaging time (see the Selecting the Square-Domain Filter and Output Low-Pass Filter section). The on-chip rms filter has a nominal corner frequency of 40 kHz. Any external capacitor acts on a 1 kΩ resistor to yield a new corner frequency for the rms filter (see Figure 1). MEASURING THE CREST FACTOR After proper calibration of the rms and envelope channels, the ratio of the two outputs gives the crest factor of the signal, when envelope output is in peak-hold mode (see the Calculation of Crest Factor (CF) section for more details). The envelope extraction that precedes rms and peak/envelope measurement is common to both channels. In addition, the rms and envelope channels share bias lines and other critical devices that are matched between the two channels, wherever possible. This ensures that the relative measurement between the two channels or the crest factor measurement of the signal is more accurate than the individual measurements of the rms value and the peak value, although these measurements in themselves are very accurate over temperature, supply, and process variations as well. Rev. A | Page 15 of 28 ADL5502 APPLICATIONS INFORMATION BASIC CONNECTIONS Figure 44 shows the basic connections for the ADL5502. The device is powered by a single supply between 2.5 V and 3.3 V, with a quiescent current of 3 mA. The VPOS pin is decoupled using 100 pF and 0.1 μF capacitors. Placing a single 75 Ω resistor at the RF input provides a broadband match of 50 Ω. More precise resistive or reactive matches can be applied for narrow frequency band use (see the RF Input Interfacing section). The rms averaging can be augmented by placing additional capacitance at CFLTR. The ac residual can be reduced further by increasing the output capacitance, COUT. The combination of the internal 100 Ω output resistance and COUT produce a low-pass filter to reduce output ripple of the VRMS output (see the Selecting the Square-Domain Filter and Output Low-Pass Filter section for more details). +VS 2.5V TO 3.3V A number of options exist for input matching. For operation at multiple frequencies, a 75 Ω shunt to ground, as shown in Figure 45, provides the best overall match. For use at a single frequency, a resistive or a reactive match can be used. By plotting the input impedance on a Smith Chart, the best value for a resistive match can be calculated. (Both input impedance and input capacitance can vary by up to ±20% around their nominal values.) Where VSWR is critical, the match can be improved with a series inductor prior to the shunt component. RF TRANSMISSION LINE DIRECTIONAL COUPLER 50Ω ATTN 75Ω RFIN ADL5502 Figure 45. Input Interfacing to Directional Coupler Resistive Tap RF Input VRMS 8 0.1µF 0.1pF CFLTR 1 ENBL FLTR VPOS VRMS PEAK 7 ROUT COUT ADL5502 2 6 VPEAK CONTROL (HIGH RESET; LOW PEAK HOLD) 07631-044 RFIN R10 75Ω 3 RFIN COMM 4 CNTL 5 Figure 44. Basic Connections for ADL5502 To measure the peak of a waveform, the control line (CNTL) must be temporally set to high (reset mode for >1 μs) and then set back to low (peak-hold mode). This allows the ADL5502 to be initialized to a known state. When setting the device to measure peak, peak-hold mode should be toggled for a period in which the input rms power and CF is not likely to change. If the ADL5502 is in peak-hold mode and the CF changes from high to low or the input power changes from high to low, a faulty peak measurement is reported. The ADL5502 simply keeps reporting the highest peak that occurred when the peakhold mode was activated and the input power or the CF was high. Unless CNTL is reset, the PEAK output does not reflect the new peak in the signal. Figure 46 shows a technique for coupling the input signal into the ADL5502 that can be applicable where the input signal is much larger than the input range of the ADL5502. A series resistor combines with the input impedance of the ADL5502 to attenuate the input signal. Because this series resistor forms a divider with the frequency dependent input impedance, the apparent gain changes greatly with frequency. However, this method has the advantage of very little power being tapped off in RF power transmission applications. If the resistor is large compared to impedance of the transmission line, the VSWR of the system is relatively unaffected. RF TRANSMISSION LINE ADL5502 Figure 46. Attenuating the Input Signal The resistive tap or series resistance, RSERIES, can be expressed as RSERIES = RIN (1 − 10ATTN/20)/(10ATTN/20) 07631-046 RSERIES RFIN 07631-045 (1) where: RIN is the input impedance of RFIN. ATTN is the desired attenuation factor in dB. RF INPUT INTERFACING The input impedance of the ADL5502 decreases with increasing frequency in both its resistive and capacitive components (see Figure 34). The resistive component varies from 330 Ω at 900 MHz to about 240 Ω at 1900 MHz. For example, if a power amplifier with a maximum output power of 28 dBm is matched to the ADL5502 input at 5 dBm, then a −23 dB attenuation factor is required. At 900 MHz, the input resistance, RIN, is 330 Ω. RSERIES = (330 Ω)(1 − 10−23/20)/(10−23/20) = 4330 Ω (2) Thus, for an attenuation of −23 dB, a series resistance of approximately 4.33 kΩ is needed. Rev. A | Page 16 of 28 ADL5502 Multiple RF Inputs Figure 47 shows a technique for combining multiple RF input signals to the ADL5502. Some applications can share a single detector for multiple bands. Three 16.5 Ω resistors in a T-network combine the three 50 Ω terminations (including the ADL5502 with the shunt 75 Ω matching component). The broadband resistive combiner ensures each port of the T-network sees a 50 Ω termination. Because there are only 6 dB of isolation from one port of the combiner to the other ports, only one band should be active at a time. BAND 1 DIRECTIONAL COUPLER 50Ω Figure 9 shows the output swings of the ADL5502 to a CW input for various supply voltages. Only at the lowest supply voltage (2.5 V) is there a reduction in the dynamic range as the input headroom decreases. VRMS Output Offset The ADL5502 has a ±1 dB error detection range of about 30 dB, as shown in Figure 10 to Figure 12 and Figure 16 to Figure 18. The error is referred to the best-fit line defined in the linear region of the output response (see the Device Calibration and Error Calculation section for more details). Below an input power of −18 dBm, the response is no longer linear and begins to lose accuracy. In addition, depending on the supply voltage, saturation may limit the detection accuracy above 12 dBm. Calibration points should be chosen in the linear region, avoiding the nonlinear ranges at the high and low extremes. BAND 2 DIRECTIONAL COUPLER 50Ω 16.5Ω 16.5Ω 75Ω RFIN 16.5Ω ADL5502 VRMS (mV) Figure 47. Combining Multiple RF Input Signals 07631-047 1k 100 LINEARITY Because the ADL5502 is a linear responding device, plots of output voltage vs. input voltage result in a straight line (see Figure 4, Figure 5, and Figure 7) and the dynamic range in dB is not clearly visible. It is more useful to plot the error on a logarithmic scale, as shown in Figure 6 and Figure 8. The deviation of the plot for the ideal straight line characteristic is caused by input stage clipping at the high end and by signal offsets at the low end. However, offsets at the low end can be either positive or negative; therefore, the linearity error vs. input level plots could also trend upwards at the low end. Figure 10, Figure 11, Figure 12, Figure 16, Figure 17, and Figure 18 show error distributions for a large population of devices at specific frequencies over temperature. It is also apparent in Figure 6 that the error at the lower portion of the dynamic range tends to shift up as frequency is increased This is due to the calibration points chosen, 0 dBm and 9 dBm (see the Device Calibration and Error Calculation section). The absolute value cell has an input impedance that varies with frequency. The result is a decrease in the actual voltage across the squaring cell as the frequency increases, reducing the conversion gain. Similarly, conversion gain is less at frequencies near 450 MHz because of the small on-chip coupling capacitor. The dynamic range is near constant over frequency, but with a decrease in conversion gain as frequency is increased. 10 –25 –20 –15 –10 –5 0 5 15 INPUT (dBm) Figure 48. VRMS Output vs. Input Level Distribution of 50 Devices, 900 MHz Frequency, Supply 3.0 V 1k PEAK (mV) 100 10 –25 –20 –15 –10 –5 0 5 10 15 INPUT (dBm) Figure 49. PEAK Output vs. Input Level Distribution of 50 Devices, 900 MHz Frequency, Supply 3.0 V Output Swing At 900 MHz, the VRMS output voltage is nominally 1.89 times the input rms voltage (a conversion gain of 1.89 V/V rms). Similarly, the PEAK output voltage is nominally 1.27 times the input rms voltage (a conversion gain of 1.27 V/V rms). The rms output voltage swings from near ground to 2.4 V on a 3.0 V supply. Figure 48 and Figure 49 show distributions of VRMS and PEAK output responses vs. the input power for multiple devices. The ADL5502 loses accuracy at low input powers as the output response begins to fanout. As the input power is reduced, the spread of the output response increases along with the error. Rev. A | Page 17 of 28 07631-049 1 –30 07631-048 1 –30 10 ADL5502 Although some devices follow the ideal linear response at very low input powers, not all devices continue the ideal linear regression to a near 0 V y-intercept. Some devices exhibit output responses that rapidly decrease and some flatten out. With no RF signal applied, the ADL5502 has a typical output offset of 15 mV (with a maximum of 100 mV) on VRMS and an offset of 14 mV (with a maximum of 100 mV) on PEAK. The square-domain filter capacitance of the ADL5502 can be augmented by connecting a capacitor between Pin 1 (FLTR) and Pin 2 (VPOS). In addition, the VRMS output of the ADL5502 can be filtered directly by placing a capacitor between VRMS (Pin 7) and ground. The PEAK output can be filtered by placing a capacitor between Pin 6 (PEAK) and ground. The combination of the on-chip, 100 Ω output series resistance and the external shunt capacitor forms a low-pass filter to reduce the residual ac. Figure 50 and Figure 51 show the effects on the residual ripple vs. the output and square-domain filter capacitor values at two communication standards with high peak-to-average ratios. Note that there is a tradeoff between ac residual and response time. Large filter capacitances increase the turn-on and pulse response times (see Figure 36, Figure 38, Figure 39, Figure 41, Figure 42, and Figure 43). Figure 52 shows the effect of the two filtering options, the output filter and the square-domain filter capacitor, on the pulse response time of the ADL5502. For more information on the effects of the filter capacitances on the response, see the Power Consumption, Enable, and PowerOn/Power-Off Response Time section. 60 OUTPUT DRIVE CAPABILITY AND BUFFERING The ADL5502 is capable of sourcing a VRMS output current of approximately 3 mA. The output current is sourced through the on-chip, 100 Ω series resistor; therefore, any load resistor forms a voltage divider with this on-chip resistance. It is recommended that the ADL5502 VRMS output drive high resistive loads to preserve output swing. If an application requires driving a low resistance load (as well as, in cases where increasing the nominal conversion gain is desired), a buffering circuit is necessary. The PEAK output is designed to drive 2 pF loads. It is recommended that the ADL5502 PEAK output drive low capacitive loads to achieve a full output response time. The effects of larger capacitive loads are particularly visible when tracking envelopes during the falling transitions. When the envelope is in a fall transition, the load capacitor discharges through the on-chip load resistance of 1.9 kΩ. If the larger capacitive load is unavoidable, the additional capacitance can be counteracted by putting a shunt resistor to ground on the PEAK output to allow for fast discharge. Such a shunt resistor also makes the ADL5502 run higher current, and it should not be reduced beyond 500 Ω. When viewing the PEAK output on an oscilloscope, a low capacitive FET probe should be used to interface with the PEAK output. This reduces the capacitance presented to the PEAK output and avoids the corresponding effects of larger capacitive loads. 50 AC RESIDUAL (V p-p) COUT 40 30 CFLTR 20 10 1 10 100 1000 CAPACITANCE (nF) SELECTING THE SQUARE-DOMAIN FILTER AND OUTPUT LOW-PASS FILTER The internal filter capacitor of the ADL5502 provides averaging in the square domain but leaves some residual ac on the output. Signals with high peak-to-average ratios, such as W-CDMA or CDMA2000, can produce ac residual levels on the ADL5502 VRMS dc output. To reduce the effects of these low frequency components in the waveforms, some additional filtering is required. Figure 50. AC Residual vs. CFLTR and COUT, W-CDMA Forward Link (4.6 dB CF) Waveform 400 350 300 AC RESIDUAL (V p-p) 250 200 150 100 50 0 COUT CFLTR 1 10 100 1000 CAPACITANCE (nF) Figure 51. AC Residual vs. CFLTR and COUT, W-CDMA Reverse Link (11.7 dB CF) Waveform Rev. A | Page 18 of 28 07631-051 07631-050 0 ADL5502 1000 900 800 CFLTR RESPONSE TIME (µs) 700 600 500 400 300 200 100 0 1 10 100 CFLTR COUT 200 180 160 140 120 100 80 60 40 20 07631-052 COUT RESPONSE TIME (µs) To improve the falling edge of the enable and pulse responses, a resistor can be placed in parallel with the output shunt capacitor. The added resistance helps to discharge the output filter capacitor. Although this method reduces the power-off time, the added load resistor also attenuates the output (see the Output Drive Capability and Buffering section). PULSED RFIN 400mV rms RF INPUT VRMS (250mV/DIV) 250mV rms 160mV rms 70mV rms 0 1000 CAPACITANCE (nF) Figure 52. Response Time vs. CFLTR and COUT The quiescent current consumption of the ADL5502 varies linearly with the size of the input signal from approximately 3 mA for no signal up to 11 mA at an input level of 0.7 V rms (10 dBm, re: 50 Ω). There is little variation in quiescent current across power supply voltage or temperature, as shown in Figure 37. The ADL5502 can be disabled either by pulling the ENBL (Pin 8) to COMM (Pin 4) or by removing the supply power to the device. Disabling the device via the ENBL function reduces the leakage current to less than 1 μA. When the device is disabled, the output impedance increases to approximately 5.5 kΩ on VRMS and 1.9 kΩ on PEAK. The turn-on time and pulse response is strongly influenced by the size of the square-domain filter and output shunt capacitor. Figure 53 shows a plot of the output response to an RF pulse on the RFIN pin, with a 0.1 μF output filter capacitor and no squaredomain filter capacitor. The falling edge is particularly dependent on the output shunt capacitance, as shown in Figure 53. PULSED RFIN VRMS 1ms/DIV Figure 54. Output Response to Various RF Input Pulse Levels, Supply 3 V, 900 MHz Frequency, Square-Domain Filter Open, Output Filter 0.1 μF with Parallel 1 kΩ The square-domain filter improves the rms accuracy for high crest factors (see the Selecting the Square-Domain Filter and Output Low-Pass Filter section), but it can hinder the response time. For optimum response time and low ac residual, both the square-domain filter and the output filter should be used. The square-domain filter at FLTR can be reduced to improve response time, and the remaining ac residual can be decreased by using the output filter, which has a smaller time constant. DEVICE CALIBRATION AND ERROR CALCULATION Because slope and intercept vary from device to device, boardlevel calibration must be performed to achieve high accuracy. In general, calibration is performed by applying two input power levels to the ADL5502 and measuring the corresponding output voltages. The calibration points are generally chosen to be within the linear operating range of the device. The best-fit line is characterized by calculating the conversion gain (or slope) and intercept using the following equations: Gain = (VVRMS2 − VVRMS1)/(VIN2 − VIN1) Intercept = VVRMS1 − (Gain × VIN1) 400mV rms RF INPUT VRMS (250mV/DIV) 250mV rms 160mV rms 07631-054 POWER CONSUMPTION, ENABLE, AND POWERON/POWER-OFF RESPONSE TIME (3) (4) 70mV rms where: VIN is the rms input voltage to RFIN. VVRMS is the voltage output at VRMS. 07631-053 VRMS 1ms/DIV Figure 53. Output Response to Various RF Input Pulse Levels, Supply 3 V, 900 MHz Frequency, Square-Domain Filter Open, Output Filter 0.1 μF Once gain and intercept are calculated, an equation can be written that allows calculation of an (unknown) input power based on the measured output voltage. VIN = (VVRMS − Intercept)/Gain (5) Rev. A | Page 19 of 28 ADL5502 For an ideal (known) input power, the law conformance error of the measured data can be calculated as ERROR (dB) = 20 × log [(VVRMS, MEASURED − Intercept)/(Gain × VIN, IDEAL)] (6) Figure 55 includes a plot of the error at 25°C, the temperature at which the ADL5502 is calibrated. Note that the error is not zero; this is because the ADL5502 does not perfectly follow the ideal linear equation, even within its operating region. The error at the calibration points is, however, equal to 0 by definition. 3 In some applications, very high accuracy is required at just one power level or over a reduced input range. For example, in a wireless transmitter, the accuracy of the high power amplifier (HPA) is most critical at or close to full power. The ADL5502 offers a tight error distribution in the high input power range, as shown in Figure 56. The high accuracy range, beginning around 4 dBm at 1900 MHz, offers 8 dB of ±0.15 dB detection error over temperature. Multiple point calibration at ambient temperature in the reduced range offers precise power measurement with near 0 dB error from −40°C to +85°C. 3 2 2 1 ERROR (dB) +85ºC +25ºC ERROR (dB) 1 +25ºC 0 +85ºC 0 –40ºC –1 –1 –40ºC –2 –2 –20 –15 –10 –5 0 5 10 15 INPUT (dBm) INPUT (dBm) Figure 55. VRMS Error from Linear Reference vs. Input at −40°C, +25°C, and +85°C vs. +25°C Linear Reference, Frequency 1900 MHz, Supply 3.0 V Figure 56. VRMS Error from +25°C Output Voltage at −40°C, +25°C, and +85°C After Ambient Normalization, Frequency 1900 MHz, Supply 3.0 V Figure 55 also includes error plots for the output voltage at −40°C and +85°C. These error plots are calculated using the gain and intercept at 25°C. This is consistent with calibration in a mass production environment where calibration at temperature is not practical. The same procedure should be followed to calculate the linearity error for the PEAK output. In this case, replace VVRMS with VPEAK in the preceding equations. Note that the high accuracy range center varies over frequency (see Figure 13, Figure 14, Figure 15, Figure 19, Figure 20, and Figure 21). CALCULATION OF CREST FACTOR (CF) The ADL5502 is a true rms power detector in combination with an envelope detector that accurately determines the crest factor of a modulated signal. The device has two outputs, VRMS and PEAK, which respectively provide the rms and envelope peak of the RF waveform present at RFIN. Therefore, these two outputs can be used to accurately calculate the crest factor of the waveform. Before CF can be measured and calculated, both of the ADL5502 outputs must be calibrated (see the Device Calibration and Error Calculation section for the calibration procedure for VRMS and PEAK). It is suggested that the calibration step be completed by applying at least two input power levels with a CW signal. The CW signal (with a CF of 0 dB) serves as the reference for the CF calculation. When the characteristics (slope and intercept) of the VRMS and PEAK outputs are known, the calibration for the CF calculation is complete. CALIBRATION FOR IMPROVED ACCURACY Another way of presenting the error function of the ADL5502 is shown in Figure 56. In this case, the dB error at hot and cold temperatures is calculated with respect to the transfer function at ambient temperature. This is a key difference in comparison to the previous plots. Up until now, the errors were calculated with respect to the ideal linear transfer function at ambient temperature. When this alternative technique is used, the error at ambient temperature becomes equal to zero by definition (see Figure 56). This plot is a useful tool for estimating temperature drift at a particular power level with respect to the (nonideal) response at ambient. The linearity and dynamic range tend to be improved artificially with this type of plot because the ADL5502 does not perfectly follow the ideal linear equation (especially outside of its linear operating range). Achieving this level of accuracy in an end application requires calibration at multiple points in the operating range of the device. Rev. A | Page 20 of 28 07631-056 –20 –15 –10 –5 0 5 10 15 07631-055 –3 –25 –3 –25 ADL5502 A three-stage process must be taken to measure and calculate the crest factor of any waveform. First, the unknown signal must be applied to the RF input and the corresponding VRMS level is measured. This level is indicated in Figure 57 as VVRMS-UNKNOWN. The RF input, VIN, is calculated using VVRMS-UNKNOWN and Equation 5. OUTPUT (V) DRIFT OVER A REDUCED TEMPERATURE RANGE Figure 59 and Figure 60 shows the error over temperature for a 1.9 GHz input signal. RMS error due to drift over temperature consistently remains within ±0.25 dB and only begins to exceed this limit when the ambient temperature rises above +65°C and drops below −30°C. For all frequencies using a reduced temperature range, higher measurement accuracy is achievable. 1.00 0.75 0.50 0.25 0 –0.25 –0.50 –0.75 +25°C –20°C +5°C +45°C +75°C –40°C –10°C +15°C +55°C +85°C –30°C 0°C +35°C +65°C PEAK OF UNKNOWN WAVEFORM VRMS OF UNKNOWN WAVEFORM (RESULT INDEPENDENT OF WAVEFORM) PEAK OF CW, CF = 0dB 1 VPEAK-UNKNOWN VVRMS-UNKNOWN VPEAK-CW 3 2 0 VIN INPUT (V rms) Figure 57. Procedure for Crest Factor Calculation Next, the CW reference level of PEAK, VPEAK-CW, is calculated using VIN (that is, the output voltage that would be seen if the incoming waveform was a CW signal). VPEAK-CW = (VIN GainPEAK) + InterceptPEAK –20 –15 –10 –5 0 5 10 15 Finally, the actual level of PEAK, VPEAK-UNKNOWN, is measured and the CF can be calculated as CF = 20·log10 (VPEAK-UNKNOWN/VPEAK-CW) INPUT (dBm) Figure 59. VRMS Typical Drift at 1.9 GHz for Various Temperatures 1.00 0.75 0.50 0.25 0 –0.25 –0.50 –0.75 +25°C –20°C +5°C +45°C +75°C –40°C –10°C +15°C +55°C +85°C –30°C 0°C +35°C +65°C (8) 10 9 8 CREST FACTOR (dB) 8-TONE WAVEFORM, 9dB CF 7 6 5 4 3 2 1 0 –1 –25 –20 4-TONE WAVEFORM, 6dB CF 07631-060 ERROR (dB) where VPEAK-CW, is used as a reference point to compare VPEAK-UNKNOWN,. If both VPEAK values are equal, then the CF is 0 dB, as shown in Figure 58 with the CW signal. Across the dynamic range, the calculated CF hovers about the 0 dB line. Likewise, for complex waveforms of 3 dB, 6 dB, and 9 dB CFs, the calculations accurately hover about the corresponding CF levels. –1.00 –25 –20 –15 –10 –5 0 5 10 15 INPUT (dBm) 2-TONE WAVEFORM, 3dB CF Figure 60. PEAK Typical Drift at 1.9 GHz for Various Temperatures CW, 0dB CF –15 –10 –5 0 5 10 15 INPUT (dBm) Figure 58. Reported Crest Factor of Various Waveforms 07631-058 Rev. A | Page 21 of 28 07631-059 (7) ERROR (dB) 07631-057 –1.00 –25 ADL5502 OPERATION AT HIGH FREQUENCIES The ADL5502 works at high frequencies but exhibits slightly higher linearity error. Figure 61 and Figure 62 show the linearity error distributions for VRMS and PEAK of 50 devices at 6000 MHz over temperature. The typical slopes at 6000 MHz are 0.87 V/V rms and 0.58 V/V rms for VRMS and PEAK, respectively. The intercepts at 6000 MHz are 0.002 V and 0.005 V for each respective output. 3 3 2 1 ERROR (dB) 0 –1 2 –2 1 ERROR (dB) –20 –15 –10 –5 0 5 10 15 0 INPUT (dBm) –1 Figure 63. VRMS Output Delta from +25°C Output Voltage for 50 Devices at −40°C and +85°C, Frequency 6000 MHz, Supply 3.0 V 3 –2 2 –20 –15 –10 –5 0 5 15 07631-061 –3 –25 10 1 ERROR (dB) INPUT (dBm) Figure 61. VRMS Output Temperature Drift from +25°C Linear Reference for 50 Devices at −40°C, +25°C, and +85°C, 6000 MHz, Supply 3.0 V 3 0 –1 2 –2 1 ERROR (dB) –20 –15 –10 –5 0 5 10 15 0 INPUT (dBm) –1 Figure 64. PEAK Output Delta from +25°C Output Voltage for 50 Devices at −40°C and +85°C, Frequency 6000 MHz, Supply 3.0 V DEVICE HANDLING –2 –3 –25 –20 –15 –10 –5 0 5 15 INPUT (dBm) Figure 62. PEAK Output Temperature Drift from +25°C Linear Reference for 50 Devices at −40°C, +25°C, and +85°C, 6000 MHz, Supply 3.0 V Due to the repeatability of the performance from part to part, compensation can be applied to reduce the effects of temperature drift and linearity error. To detect larger dynamic ranges at higher frequencies, the transfer function at ambient temperature can be calibrated, thus eliminating the linearity error. This technique is discussed in detail in the Calibration for Improved Accuracy section. Figure 63 and Figure 64 show the temperature drift distributions for both outputs of 50 devices at 6000 MHz. 07631-062 10 The wafer-level chip scale package consists of solder bumps connected to the active side of the die. The part is lead-free with 95.5% tin, 4.0% silver, and 0.5% copper solder bump composition. The WLCSP can be mounted on printed circuit boards using standard surface-mount assembly techniques; however, caution should be taken to avoid damaging the die. See the AN-617 Application Note, MicroCSP Wafer Level Chip Scale Package, for additional information. WLCSP devices are bumped die; therefore, the exposed die can be sensitive to light, which can influence specified limits. Lighting in excess of 600 lux can degrade performance. Rev. A | Page 22 of 28 07631-064 –3 –25 07631-063 –3 –25 ADL5502 EVALUATION BOARD Figure 65 shows the schematic of the ADL5502 evaluation board. The board is powered by a single supply in the 2.5 V to 3.3 V range. The power supply is decoupled by 100 pF and 0.1 μF capacitors. Table 4 details the various configuration options of the evaluation board. Figure 66 and Figure 67 show the component and circuit layouts of the evaluation board. The RF input has a broadband match of 50 Ω using a single 75 Ω resistor at R10. More precise matching at spot frequencies is possible using the pads for C15, C16, and R10 (see the RF Input Interfacing section). The two outputs, accessible via the SMAs labeled VRMS and VENV, provide the rms response and the envelope/peak-hold measurement of the RF input power level. The device must be enabled by switching SW1 to high (setting the switch to the position opposite that of the SW1 label). (P1 – 8) R2 0Ω ENBL 8 1 The device is placed in peak-hold mode by placing the switch SW2 in the position closes to the SW2 label. Envelope-tracking mode is possible by setting SW2 in the opposite switch position (away from the SW2 label). A signal generator can drive the control mode via the SMA labeled CNTL (see Table 4 for more details). Operating in Peak-Hold Mode To operate the device in peak-hold mode, the control line must be temporally set to high (reset mode for >1 μs) and then set back to low (peak-hold mode). This allows the ADL5502 initialize to a known state. Land Pattern and Soldering Information Pad diameters of 0.28 mm are recommended with a solder paste mask opening of 0.38 mm. For the RF input trace, a trace width of 0.30 mm is used, which corresponds to a 50 Ω characteristic impedance for the dielectric material being used (FR4). All traces going to the pads are tapered down to 0.15 mm. For the RFIN line, the length of the tapered section is 0.20 mm. (P1 – 4) R3 0Ω C18 (OPEN) R11 (OPEN) (P1 – 6) R4 0Ω VRMS SW1 VPOS R1 0Ω FLTR VPOS RFIN ENBL VRMS 7 PEAK 6 CNTL 5 R13 0Ω R6 0Ω C19 (OPEN) R12 (OPEN) (P1 – 6) R9 0Ω CNTL R5 0Ω VENV C17 10nF VPOS 2 ADL5502 C13 0.1µF C14 100pF 3 COMM 4 RFIN C15 R10 75Ω C16 VPOS (P1 – 1) C12 100pF C11 0.1µF C20 (OPEN) R8 10kΩ SW2 R7 10kΩ VPOS Figure 65. Evaluation Board Schematic 07631-065 07631-066 Figure 66. Layout of Evaluation Board, Component Side Rev. A | Page 23 of 28 Figure 67. Layout of Evaluation Board, Circuit Side 07631-067 ADL5502 Table 4. Evaluation Board Configuration Options Component VPOS, GND C13, C14 C17 R10, C15, C16 Description Ground and Supply Vector Pins. Power Supply Decoupling. Nominal supply decoupling of 0.01 μF and 100 pF. Filter Capacitor. The internal rms averaging capacitor can be augmented by placing additional capacitance in C17. RF Input interface. The 75 Ω resistor at R10 combines with the ADL5502 internal input impedance to give a broadband input impedance of around 50 Ω. The pads for components C15, C16, and R10 can be used for more precise matching at a particular frequency. Output Filtering. The combination of the internal 100 Ω output resistance and C18 produce a low-pass filter to reduce output ripple of the VRMS output. Similarly, C19 and the internal 100 Ω output resistance form a low-pass filter at the PEAK output. Either output can be scaled down using the resistor divider pads, R3, R11, R6, and R12. Device Enable. When the switch is set toward the SW1 label, the ENBL pin is grounded (through the 0 Ω resistor) putting the device in power-down mode. In the opposite switch position, the ENBL pin is connected to VPOS and the ADL5502 is in enable mode. While the switch is in the disabled position, the ENBL pin can be driven by a signal generator via the SMA labeled ENBL. In this case, R1 must be removed or changed to provide a 50 Ω match. Control Interface. When the switch is set toward the SW2 label, the CNTL pin is grounded (through a 10 kΩ resistor) putting the device in peak-hold mode. In the opposite switch position, the pin is connected to VPOS (through a 10 kΩ resistor) and the ADL5502 is in reset mode. While the switch is in the peak-hold position, the CNTL pin can be driven by a signal generator via the SMA labeled CNTL. In this case, R8 may be removed or changed to provide a 50 Ω match. R13 and C20 allow for a low-pass filter design for the control pin. Alternate Interface. The end connector, P1, allows access to various ADL5502 signals. These signal paths are only used during factory test and characterization. Default Condition Not applicable C13 = 0.1 μF (Size 0402) C14 = 100 pF (Size 0402) C17 = 10 nF (Size 0402) R10 = 75 Ω (Size 0402) C15, C16 = 0 Ω (Size 0402) R3, R6 = 0 Ω (Size 0402) R11, R12 = open (Size 0402) C18, C19 = open (Size 0402) R1 = 0 Ω (Size 0402) SW1 = away from SW1 label R3, R6, R11, R12, C18, C19 R1, SW1 R7, R8, R13, C20, SW2 R7, R8 = 10 kΩ (Size 0402) R13 = 0 Ω (Size 0402) C20 = Open (Size 0402) SW2 = away from SW1 label R2, R4, R5, R9, C11, C12 R2, R4, R5, R9 = 0 Ω (Size 0402) C11 = 0.1 μF (Size 0402) C12 = 100 pF (Size 0402) Rev. A | Page 24 of 28 ADL5502 OUTLINE DIMENSIONS 1.500 1.460 SQ 1.420 0.625 0.570 0.514 SEATING PLANE 0.345 0.295 0.245 3 2 1 A BALL A1 IDENTIFIER 0.50 BALL PITCH B C TOP VIEW Figure 68. 8-Ball Wafer Level Chip Scale Package [WLCSP] (CB-8-3) Dimensions shown in millimeters ORDERING GUIDE Model 1 ADL5502ACBZ-P7 ADL5502ACBZ-P2 ADL5502-EVALZ 1 Temperature Range –40°C to +85°C –40°C to +85°C Package Description 8-Ball WLCSP, 7” Pocket Tape and Reel 8-Ball WLCSP, 7” Pocket Tape and Reel Evaluation Board Package Option CB-8-3 CB-8-3 Branding Q1C Q1C 03-17-2009-A 0.355 0.330 0.304 0.05 NOM COPLANARITY 0.270 0.240 0.210 BOTTOM VIEW (BALL SIDE UP) Ordering Quantity 3,000 250 Z = RoHS Compliant Part. Rev. A | Page 25 of 28 ADL5502 NOTES Rev. A | Page 26 of 28 ADL5502 NOTES Rev. A | Page 27 of 28 ADL5502 NOTES ©2008–2011 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D07631-0-1/11(A) Rev. A | Page 28 of 28