OPA3681
OPA3
681
OPA3
681
www.ti.com
Triple Wideband, Current-Feedback
OPERATIONAL AMPLIFIER With Disable
TM
FEATURES
DESCRIPTION
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The OPA3681 sets a new level of performance for broadband
triple current-feedback op amps. Operating on a very low
6mA/ch supply current, the OPA3681 offers a slew rate and
output power normally associated with a much higher supply
current. A new output stage architecture delivers a high output
current with minimal voltage headroom and crossover distortion. This gives exceptional single-supply operation. Using a
single +5V supply, the OPA3681 can deliver a 1V to 4V output
swing with over 100mA drive current and 150MHz bandwidth.
This combination of features makes the OPA3681 an ideal RGB
line driver or single-supply ADC input driver.
WIDEBAND +5V OPERATION: 225MHz (G = +2)
UNITY GAIN STABLE: 280MHz (G = 1)
HIGH OUTPUT CURRENT: 150mA
OUTPUT VOLTAGE SWING: ±4.0V
HIGH SLEW RATE: 2100V/µs
LOW SUPPLY CURRENT: 6mA/ch
LOW DISABLED CURRENT: 300µA/ch
IMPROVED HIGH FREQUENCY PINOUT
APPLICATIONS
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The OPA3681’s low 6mA/ch supply current is precisely trimmed
at 25°C. This trim, along with low drift over temperature,
guarantees lower guaranteed maximum supply current than
competing products. System power may be further reduced by
using the optional disable control pin. Leaving this disable pin
open, or holding it high, gives normal operation. If pulled low,
the OPA3681 supply current drops to less than 300µA/ch while
the output goes into a high impedance state. This feature may be
used for power savings or for video MUX applications.
RGB AMPLIFIERS
WIDEBAND INA
BROADBAND VIDEO BUFFERS
HIGH SPEED IMAGING CHANNELS
PORTABLE INSTRUMENTS
ADC BUFFERS
ACTIVE FILTERS
CABLE DRIVERS
OPA3681 RELATED PRODUCTS
Voltage Feedback
Current Feedback
Fixed Gain
+5
1/3
OPA3681
–5
250Ω
1/3
OPA3681
250Ω
10 (V1 – V2)
–5
499Ω
301Ω
23
f–3dB
17
14
11
499Ω
1/3
OPA3681
–5
OPA3680
OPA3681
OPA3682
20
66.5Ω
V2
TRIPLES
OPA2680
OPA2681
OPA2682
26
+5
301Ω
+5
DUALS
OPA680
OPA681
OPA682
DIFFERENTIAL TO SINGLE-ENDED
FREQUENCY RESPONSE
Gain (dB)
V1
SINGLES
8
5
0.1M
High Speed INA (>120MHz)
1M
10M
100M
1G
Frequency (Hz)
Copyright © 1999, Texas Instruments Incorporated
SBOS095A
Printed in U.S.A. January, 2001
SPECIFICATIONS: VS = ±5V
RF = 499Ω, RL = 100Ω, and G = +2, (Figure 1 for AC performance only), unless otherwise noted.
OPA3681E, U
TYP
CONDITIONS
+25°C
G = +1, RF = 549Ω
G = +2, RF = 499Ω
G = +5, RF = 365Ω
G = +10, RF = 182Ω
G = +2, VO = 0.5Vp-p
RF = 453, VO = 0.5Vp-p
G = +2, VO = 5Vp-p
G = +2, 4V Step
G = +2, VO = 0.5V Step
G = +2, 5V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 5MHz, VO = 2Vp-p
RL = 100Ω
RL ≥ 500Ω
RL = 100Ω
RL ≥ 500Ω
f > 1MHz
f > 1MHz
f > 1MHz
G = +2, NTSC, VO = 1.4Vp, RL = 150Ω
RL = 37.5Ω
G = +2, NTSC, VO = 1.4Vp, RL = 150Ω
RL = 37.5Ω
Input Referred, f = 5MHz, All Hostile
280
220
185
125
90
0.4
150
2100
1.7
2.0
12
8
PARAMETER
AC PERFORMANCE (Figure 1)
Small Signal Bandwidth (VO = 0.5Vp-p)
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large Signal Bandwidth
Slew Rate
Rise/Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd Harmonic
3rd Harmonic
Input Voltage Noise
Non-Inverting Input Current Noise
Inverting Input Current Noise
Differential Gain
Differential Phase
Crosstalk
DC PERFORMANCE(4)
Open-Loop Transimpedance Gain (ZOL)
Input Offset Voltage
Average Offset Voltage Drift
Non-Inverting Input Bias Current
Average Non-Inverting Input Bias Current Drift
Inverting Input Bias Current
Average Inverting Input Bias Current Drift
INPUT
Common-Mode Input Range(5)
Common-Mode Rejection (CMR)
Non-Inverting Input Impedance
Inverting Input Resistance (RI )
OUTPUT
Voltage Output Swing
Current Output, Sourcing
Current Output, Sinking
Closed-Loop Output Impedance
DISABLE (Disabled Low)
Power Down Supply Current (+VS)
Disable Time
Enable Time
Off Isolation
Output Capacitance in Disable
Turn On Glitch
Turn Off Glitch
Enable Voltage
Disable Voltage
Control Pin Input Bias Current (DIS)
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage Range
Max Quiescent Current (3 Channels)
Min Quiescent Current (3 Channels)
Power Supply Rejection Ratio (–PSRR)
TEMPERATURE RANGE
Specification: E, U
Thermal Resistance, θJA
E SSOP-16
U SO-16
–75
–81
–80
–95
2.2
12
15
0.001
0.005
0.01
0.05
–55
GUARANTEED
+25°C(2)
0°C to
70°C(3)
–40°C to
+85°C(3)
220
210
190
50
2
45
4
45
1600
1600
1200
UNITS
MIN/ TEST
MAX LEVEL(1)
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
ns
typ
min
typ
typ
min
max
typ
min
typ
typ
typ
typ
C
B
C
C
B
B
C
B
C
C
C
C
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
pA/√Hz
%
%
deg
deg
dBc
typ
typ
typ
typ
max
max
max
typ
typ
typ
typ
typ
C
C
C
C
B
B
B
C
C
C
C
C
3.0
14
18
3.4
15
18
3.6
15
19
100
±1.3
56
±5
+30
+55
±10
±40
56
±6.5
+35
±65
–400
±50
–125
56
±7.5
+40
±85
–450
±55
–150
kΩ
mV
µV/°C
µA
nA/°C
µA
nA°/C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
±3.4
±3.3
46
±3.2
45
Open Loop
±3.5
52
100 || 2
42
V
dB
kΩ || pF
Ω
min
min
typ
typ
A
A
C
C
No Load
RL = 100Ω
VO = 0
VO = 0
G = +2, f = 100kHz
±4.0
±3.9
+190
–150
0.03
±3.8
±3.7
±3.7
±3.6
+140
–130
±3.6
±3.3
+80
–80
V
V
mA
mA
Ω
min
min
min
min
typ
A
A
A
A
C
VDIS = 0, All Channels
–960
100
25
70
4
±50
±20
3.3
1.8
100
typ
typ
typ
typ
typ
typ
typ
min
max
max
C
C
C
C
C
C
C
A
A
A
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
G = +2, 5MHz
G = +2, RL = 150Ω, VIN = 0
G = +2, RL = 150Ω, VIN = 0
VDIS = 0, Each Channel
+160
–135
3.5
1.7
160
3.6
1.6
160
3.7
1.5
160
µA
ns
ns
dB
pF
mV
mV
V
V
µA
±6
±6
19.5
16.5
50
±6
19.8
15.0
49
V
V
mA
mA
dB
typ
max
max
min
min
C
A
A
A
A
–40 to +85
°C
typ
C
100
100
°C/W
°C/W
typ
typ
C
C
±5
VS = ±5V
VS = ±5V
Input Referred
47
18
18
58
19.2
16.8
52
NOTES: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.
(C) Typical value only for information. (2) Junction temperature = ambient for 25°C guaranteed specifications. (3) Junction temperature = ambient at low temperature
limit: Junction temperature = ambient +23°C at high temperature limit for over temperature guaranteed specifications. (4) Current is considered positive out of node.
VCM is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ± CMIR limits.
2
OPA3681
SBOS095A
SPECIFICATIONS: VS = +5V
RF = 499Ω, RL = 100Ω to VS /2, G = +2, (Figure 2 for AC performance only), unless otherwise noted.
OPA3681E, U
TYP
PARAMETER
AC PERFORMANCE (Figure 2)
Small Signal Bandwidth (VO = 0.5Vp-p)
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large Signal Bandwidth
Slew Rate
Rise/Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd Harmonic
3rd Harmonic
Input Voltage Noise
Non-Inverting Input Current Noise
Inverting Input Current Noise
DC PERFORMANCE(4)
Open-Loop Transimpedance Gain (ZOL)
Input Offset Voltage
Average Offset Voltage Drift
Non-Inverting Input Bias Current
Average Non-Inverting Input Bias Current Drift
Inverting Input Bias Current
Average Inverting Input Bias Current Drift
INPUT
Least Positive Input Voltage(5)
Most Positive Input Voltage(5)
Common-Mode Rejection (CMR)
Non-Inverting Input Impedance
Inverting Input Resistance (RI )
OUTPUT
Most Positive Output Voltage
Least Positive Output Voltage
Current Output, Sourcing
Current Output, Sinking
Closed-Loop Output Impedance
DISABLE (Disable Low)
Power Down Supply Current (+VS)
Disable Time
Enable Time
Off Isolation
Output Capacitance in Disable
Turn On Glitch
Turn Off Glitch
Enable Voltage
Disable Voltage
Control Pin Input Bias Current (DIS)
POWER SUPPLY
Specified Single Supply Operating Voltage
Maximum Single Supply Operating Voltage
Max Quiescent Current (3 Channels)
Min Quiescent Current (3 Channels)
Power Supply Rejection Ratio (+PSRR)
TEMPERATURE RANGE
Specification: E, U
Thermal Resistance, θJA
E SSOP-16
U SO-16
GUARANTEED
+25°C(2)
0°C to
70°C(3)
–40°C to
+85°C(3)
180
140
110
50
2
35
4
23
700
680
570
CONDITIONS
+25°C
G = +1, RF = 549Ω
G = +2, RF = 499Ω
G = +5, RF = 365Ω
G = +10, RF = 182Ω
G = +2, VO < 0.5Vp-p
RF = 649Ω, VO < 0.5Vp-p
G = +2, VO = 2Vp-p
G = +2, 2V Step
G = +2, VO = 0.5V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 5MHz, VO = 2Vp-p
RL = 100Ω to VS /2
RL ≥ 500Ω to VS /2
RL = 100Ω to VS /2
RL ≥ 500Ω to VS /2
f > 1MHz
f > 1MHz
f > 1MHz
250
225
180
165
100
0.4
200
830
1.5
2.0
14
9
–75
–79
–68
–70
2.2
12
15
3
14
18
3.4
14
18
VO = VS /2, RL = 100Ω to VS /2
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
100
±1
±5
60
+40
+65
±5
±20
VCM = VS/2
Open Loop
1.5
3.5
51
100 || 2
44
No Load
RL = 100Ω, 2.5V
No Load
RL = 100Ω, 2.5V
VO = VS /2
VO = VS /2
G = +2, f = 100kHz
4
3.9
1
1.1
150
–110
0.03
VDIS = 0, All Channels
–810
100
25
65
4
±50
±20
3.3
1.8
100
G = +2, 5MHz
G = +2, RL = 150Ω, VIN = VS /2
G = +2, RL = 150Ω, VIN = VS /2
VDIS = 0, Each Channel
MIN/ TEST
MAX LEVEL(1)
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
ns
typ
min
typ
typ
min
max
typ
min
typ
typ
typ
typ
C
B
C
C
B
B
C
B
C
C
C
C
3.6
15
19
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
pA/√Hz
typ
typ
typ
typ
max
max
max
C
C
C
C
B
B
B
53
±6.0
+15
+75
–300
±25
–125
51
±7
+20
+95
–350
±35
–175
kΩ
mV
µV/°C
µA
nA/°C
µA
nA /°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
1.6
3.4
45
1.7
3.3
44
1.8
3.2
44
V
V
dB
kΩ || pF
Ω
max
min
min
typ
typ
A
A
A
C
C
3.8
3.7
1.2
1.3
110
–75
3.7
3.6
1.3
1.4
110
–70
3.5
3.4
1.5
1.6
60
–50
V
V
V
V
mA
mA
Ω
min
min
max
max
min
min
typ
A
A
A
A
A
A
C
µA
ns
ns
dB
pF
mV
mV
V
V
µA
typ
typ
typ
typ
typ
typ
typ
min
max
typ
C
C
C
C
C
C
C
A
A
C
V
V
mA
mA
dB
typ
max
max
min
typ
C
A
A
A
C
–40 to +85
°C
typ
C
100
100
°C/W
°C/W
typ
typ
C
C
3.5
1.7
3.6
1.6
3.7
1.5
12
16.2
12.3
12
16.5
11.1
12
16.5
10.8
5
VS = +5V
VS = +5V
Input Referred
UNITS
14.4
14.4
48
NOTES: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.
(C) Typical value only for information. (2) Junction temperature = ambient for 25°C guaranteed specifications. (3) Junction temperature = ambient at low temperature
limit: Junction temperature = ambient +23°C at high temperature limit for over temperature guaranteed specifications. (4) Current is considered positive out of node.
VCM is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ±CMIR limits.
OPA3681
SBOS095A
3
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
Power Supply .............................................................................. ±6.5VDC
Internal Power Dissipation(1) ............................ See Thermal Information
Differential Input Voltage .................................................................. ±1.2V
Input Voltage Range ............................................................................ ±VS
Storage Temperature Range: E, U ................................ –40°C to +125°C
Lead Temperature (soldering, 10s) .............................................. +300°C
Junction Temperature (TJ ) ........................................................... +175°C
Top View
SSOP-16 , SO-16
OPA3681
NOTE:: (1) Packages must be derated based on specified θJA. Maximum TJ
must be observed.
ELECTROSTATIC
DISCHARGE SENSITIVITY
Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. Burr-Brown Corporation recommends that all integrated circuits be handled and stored
using appropriate ESD protection methods.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes
could cause the device not to meet published specifications.
–IN A
1
16 DIS A
+IN A
2
15 +VS
DIS B
3
14 OUT A
–IN B
4
13 –VS
+IN B
5
12 OUT B
DIS C
6
11 +VS
–IN C
7
10 OUT C
+IN C
8
9
–VS
PACKAGE/ORDERING INFORMATION
PRODUCT
PACKAGE
PACKAGE
DRAWING
NUMBER(1)
OPA3681E
SSOP-16 Surface Mount
322
–40°C to +85°C
OPA3681E
"
"
"
"
SO-16 Surface Mount
265
–40°C to +85°C
OPA3681U
"
"
"
"
"
OPA3681U
"
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER(2)
TRANSPORT
MEDIA
OPA3681E/250
OPA3681E/2K5
OPA3681U
OPA3681U/2K5
Tape and Reel
Tape and Reel
Rails
Tape and Reel
NOTES: (1) For detailed drawing and dimension table, please see end of data sheet. (2) Models with a slash (/) are available only in Tape and Reel in the quantities
indicated (e.g., /2K5 indicates 2500 devices per reel). Ordering 2500 pieces of “OPA3681E/2K5” will get a single 2500-piece Tape and Reel.
4
OPA3681
SBOS095A
TYPICAL PERFORMANCE CURVES: VS = ±5V
G = +2, RF = 499Ω, and RL = 100Ω, unless otherwise noted (see Figure 1).
SMALL-SIGNAL FREQUENCY RESPONSE
LARGE-SIGNAL FREQUENCY RESPONSE
2
8
G = +1, RF = 549Ω
G = +2, RL = 100Ω
7
G = +2, RF = 499Ω
0
6
–1
Gain (1dB/div)
Normalized Gain (1dB/div)
1
VO = 0.5Vp-p
–2
–3
–4
G = +5, RF = 365Ω
–5
–6
5
2Vp-p
4
3
2
7Vp-p
0
G = +10, RF = 182Ω
–7
–1
–8
–2
0
125MHz
250MHz
0
125MHz
Frequency (25MHz/div)
SMALL-SIGNAL PULSE RESPONSE
LARGE-SIGNAL PULSE RESPONSE
+4
G = +2
VO = 5Vp-p
+3
Output Voltage (1V/div)
Output Voltage (100mV/div)
G = +2
VO = 0.5Vp-p
300
200
100
0
–100
–200
+2
+1
0
–1
–2
–3
–300
–4
–400
Time (5ns/div)
Time (5ns/div)
ALL HOSTILE CROSSTALK
LARGE-SIGNAL DISABLE/ENABLE RESPONSE
4.0
VDIS
2.0
2.0
0
0
Output Voltage
2.0
1.6
1.2
0.8
0
4.0
–20
–30
–40
Crosstalk (dB)
6.0
VDIS (2V/div)
5.0
0.4
250MHz
Frequency (25MHz/div)
400
Output Voltage (400mV/div)
1Vp-p
4Vp-p
1
–50
–60
–70
–80
G = +2
VIN = +1V
–90
–100
Time (50ns/div)
0.3
1
10
100
300
Frequency (MHz)
OPA3681
SBOS095A
5
TYPICAL PERFORMANCE CURVES: VS = ±5V
(Cont.)
G = +2, RF = 499Ω, and RL = 100Ω, unless otherwise noted (see Figure 1).
5MHz 3rd HARMONIC DISTORTION
vs OUTPUT VOLTAGE
5MHz 2nd HARMONIC DISTORTION
vs OUTPUT VOLTAGE
–50
–65
3rd Harmonic Distortion (dBc)
2nd Harmonic Distortion (dBc)
–60
RL = 100Ω
–70
RL = 200Ω
–75
–80
RL = 500Ω
–85
0.1
1
0.1
1
5
Output Voltage Swing (Vp-p)
Output Voltage (Vp-p)
10MHz 2nd HARMONIC DISTORTION
vs OUTPUT VOLTAGE
10MHz 3rd HARMONIC DISTORTION
vs OUTPUT VOLTAGE
–50
3rd Harmonic Distortion (dBc)
2nd Harmonic Distortion (dBc)
RL = 200Ω
RL = 100Ω
–80
10
–50
RL = 500Ω
–60
RL = 100Ω
RL = 200Ω
–70
–80
–90
–60
RL = 200Ω
RL = 500Ω
–70
RL = 100Ω
–80
–90
0.1
1
5
0.1
1
5
Output Voltage (Vp-p)
Output Voltage (Vp-p)
20MHz 2nd HARMONIC DISTORTION
vs OUTPUT VOLTAGE
20MHz 3rd HARMONIC DISTORTION
vs OUTPUT VOLTAGE
–50
–50
3rd Harmonic Distortion (dBc)
2nd Harmonic Distortion (dBc)
RL = 500Ω
–70
–90
–90
RL = 500Ω
–60
RL = 200Ω
RL = 100Ω
–70
–80
–90
RL = 200Ω
RL = 500Ω
–60
RL = 100Ω
–70
–80
–90
0.1
1
Output Voltage (Vp-p)
6
–60
5
0.1
1
5
Output Voltage (Vp-p)
OPA3681
SBOS095A
TYPICAL PERFORMANCE CURVES: VS = ±5V
(Cont.)
G = +2, RF = 499Ω, and RL = 100Ω, unless otherwise noted (see Figure 1).
3rd HARMONIC DISTORTION
vs FREQUENCY
2nd HARMONIC DISTORTION
vs FREQUENCY
–50
VO = 2Vp-p
RL = 100Ω
–50
G = +10, RF = 180Ω
G = +5, RF = 261Ω
–60
–70
G = +2, RF = 402Ω
–80
3rd Harmonic Distortion (dBc)
2nd Harmonic Distortion (dBc)
–40
0.1
1
10
G = +2, RF = 499Ω
–80
0.1
1
10
Frequency (MHz)
Frequency (MHz)
INPUT VOLTAGE AND CURRENT NOISE DENSITY
TWO-TONE, 3RD-ORDER
INTERMODULATION SPURIOUS
20
–40
Inverting Input Current Noise
15.1pA/√Hz
10
Non-Inverting Input Current Noise
12.2pA/√Hz
2.2nV/√Hz
Voltage Noise
3rd-Order Spurious Level (dBc)
Current Noise (pA/√Hz)
G = +5, RF = 365Ω
–70
20
100
dBc = dB below carriers
–45
–50
50MHz
–55
–60
–65
–70
20MHz
–75
–80
10MHz
–85
Load Power at Matched 50Ω Load
–90
1
100
1k
10k
100k
1M
10M
–8
–6
Frequency (Hz)
–4
–2
0
2
4
6
8
10
Single-Tone Load Power (dBm)
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
60
Gain to Capacitive Load (3dB/div)
15
50
40
RS (Ω)
G = +10, RF = 182Ω
–60
–90
–90
Voltage Noise (nV/√Hz)
VO = 2Vp-p
RL = 100Ω
30
20
10
12
CL = 10pF
9
CL = 22pF
6
3
CL = 47pF
0
–3
VIN
RS
VO
–6
499Ω
–9
CL
1kΩ
499Ω
–12
1kΩ is optional.
0
–15
1
10
Capacitive Load (pF)
OPA3681
SBOS095A
100
0
CL = 100pF
150MHz
300MHz
Frequency (30MHz/div)
7
TYPICAL PERFORMANCE CURVES: VS = ±5V
(Cont.)
G = +2, RF = 499Ω, and RL = 100Ω, unless otherwise noted (see Figure 1).
OPEN-LOOP TRANSIMPEDANCE GAIN/PHASE
120
55
CMR
50
45
40
35
30
25
100
–40
| ZOL|
80
–80
60
–120
40
–160
20
–200
0
20
102
103
104
105
106
107
104
108
105
106
COMPOSITE VIDEO dG/dP
TYPICAL DC DRIFT OVER TEMPERATURE
5
0.05
Positive Video
Negative Sync
Input Offset Voltage (mV)
dP
dG/dP (%/°)
50
4
0.04
0.03
0.02
0.01
dG
1
30
2
20
Inverting
1
0
2
3
0
–1
–10
–2
–20
–3
–30
–4
–40
–50
–40
4
–20
0
40
60
80
100
120
140
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
10
200
Output Current Limited
Sourcing Output Current
Sinking Output Current
1W Internal
Power Limit
1-Channel
Only
25Ω
Load Line
0
50Ω Load Line
–1
100Ω Load Line
–2
Supply Current (mA)
VO (Volts)
20
Ambient Temperature (°C)
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
5
1
10
VIO
Number of 150Ω Loads
2
40
Non-Inverting Input Bias Current
3
–5
0
3
–240
109
108
Frequency (Hz)
Frequency (Hz)
4
107
7.5
150
5
100
Quiescent Supply Current
2.5
50
–3
–4
1W Internal
Power Limit
Output Current Limit
0
–5
–300
–200
–100
0
IO (mA)
8
100
Input Bias Currents (µA)
–PSR
0
∠ ZOL
200
300
0
–40
–20
0
20
40
60
80
100
120
140
Ambient Temperature (°C)
OPA3681
SBOS095A
Output Current (mA)
60
Transimpedance Gain (20dBΩ/div)
Rejection Ratio (dB)
65
+PSR
Transimpedance Phase (40°/div)
CMR AND PSR vs FREQUENCY
70
TYPICAL PERFORMANCE CURVES: VS = +5V
G = +2, RF = 499Ω, and RL = 100Ω, unless otherwise noted (see Figure 1).
SMALL-SIGNAL FREQUENCY RESPONSE
2
VO = 0.5Vp-p
G = +2,
RF = 499Ω
0
–1
G = +1,
RF = 549Ω
–2
–3
G = +5,
RF = 365Ω
–4
–5
VO = 1Vp-p
5
4
VO = 2Vp-p
3
2
1
0
–7
–1
–8
–2
0
125
250
0
125
Frequency (25MHz/div)
SMALL-SIGNAL PULSE RESPONSE
LARGE-SIGNAL PULSE RESPONSE
4.5
2.8
2.7
2.6
2.5
2.4
2.3
2.2
G = +2
VO = 2Vp-p
4.1
Output Voltage (400mV/div)
G = +2
VO = 0.5Vp-p
2.9
Output Voltage (100mV/div)
250
Frequency (25MHz/div)
2.10
3.7
3.3
2.9
2.5
2.1
1.7
1.3
0.9
2.1
0.5
2.0
Time (5ns/div)
Time (5ns/div)
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
70
Gain to Capacitive Load (3dB/div)
15
60
50
RS (Ω)
VO = 0.5Vp-p
6
G = +10,
RF = 182Ω
–6
G = +2
RL = 100Ω to 2.5V
7
Gain (1dB/div)
1
Normalized Gain (1dB/div)
LARGE-SIGNAL FREQUENCY RESPONSE
8
40
30
20
10
0
CL = 10pF
CL = 47pF
12
9
CL = 22pF
6
+5V
3
0
0.1µF
806Ω
VI
–3
0.1µF
57.6Ω
VO
806Ω
RS C
L
–6
1kΩ
499Ω
–9
499Ω
–12
CL = 100pF
0.1µF
–15
1
10
Capacitive Load (pF)
OPA3681
SBOS095A
100
0
100MHz
200MHz
Frequency (20MHz/div)
9
TYPICAL PERFORMANCE CURVES: VS = +5V
(Cont.)
G = +2, RF = 499Ω, and RL = 100Ω, unless otherwise noted (see Figure 1).
2nd HARMONIC DISTORTION
vs FREQUENCY
3rd HARMONIC DISTORTION
vs FREQUENCY
–50
VO = 2Vp-p
RL = 100Ω
G = +10, RF = 182Ω
3rd Harmonic Distortion (dBc)
2nd Harmonic Distortion (dBc)
–50
–60
G = +5, RF = 365Ω
–70
–80
G = +2, RF = 499Ω
–90
G = +10, RF = 182Ω
G = +5, RF = 365Ω
–60
–70
G = +2, RF = 499Ω
–80
–90
0.1
1
10
20
0.1
1
Frequency (MHz)
Frequency (MHz)
2nd HARMONIC DISTORTION
vs FREQUENCY
3rd HARMONIC DISTORTION
vs FREQUENCY
–50
RL = 500Ω
–60
RL = 200Ω
–70
RL = 100Ω
–80
–90
20
10
20
VO = 2Vp-p
G = +2
–60
RL = 200Ω
–70
RL = 100Ω
–80
RL = 500Ω
–90
0.1
1
10
20
0.1
1
Frequency (MHz)
Frequency (MHz)
CLOSED-LOOP OUTPUT IMPEDANCE
TWO-TONE, 3rd-ORDER SPURIOUS LEVEL
–45
10
dBc = dB Below Carrier
–50
+5
50MHz
Output Impedance (Ω)
3rd-Order Spurious (dBc)
10
–50
VO = 2Vp-p
G = +2
3rd Harmonic Distortion (dBc)
2nd Harmonic Distortion (dBc)
VO = 2Vp-p
RL = 100Ω
–55
–60
–65
20MHz
–70
10MHz
–75
50Ω
1
1/3
OPA3681
ZO
499Ω
499Ω
0.1
–5
–80
Load Power at Matched 50Ω Load
0.01
–85
–14
–12
–10
–8
–6
–4
–2
Single-Tone Load Power (dBm)
10
0
2
10k
100k
1M
10M
100M
Frequency (Hz)
OPA3681
SBOS095A
APPLICATIONS INFORMATION
WIDEBAND CURRENT-FEEDBACK OPERATION
The OPA3681 gives the exceptional AC performance of a
wideband current-feedback op amp with a highly linear,
high power output stage. Requiring only 6mA/ch quiescent
current, the OPA3681 will swing to within 1V of either
supply rail and deliver in excess of 135mA guaranteed at
room temperature. This low output headroom requirement,
along with supply voltage independent biasing, gives remarkable single (+5V) supply operation. The OPA3681 will
deliver greater than 200MHz bandwidth driving a 2Vp-p
output into 100Ω on a single +5V supply. Previous boosted
output stage amplifiers have typically suffered from very
poor crossover distortion as the output current goes through
zero. The OPA3681 achieves a comparable power gain with
much better linearity. The primary advantage of a currentfeedback op amp over a voltage-feedback op amp is that AC
performance (bandwidth and distortion) is relatively independent of signal gain.
Figure 1 shows the DC-coupled, gain of +2, dual power
supply circuit configuration used as the basis of the ±5V
Specifications and Typical Performance Curves. For test
purposes, the input impedance is set to 50Ω with a resistor
to ground and the output impedance is set to 50Ω with a
series output resistor. Voltage swings reported in the specifications are taken directly at the input and output pins while
load powers (dBm) are defined at a matched 50Ω load. For
the circuit of Figure 1, the total effective load will be 100Ω
|| 998Ω. The disable control line (DIS) is typically left open
to guarantee normal amplifier operation. One optional component is included in Figure 1. In addition to the usual power
supply de-coupling capacitors to ground, a 0.1µF capacitor
0.1µF
+5V
+VS
is included between the two power supply pins. In practical
PC board layouts, this optionally added capacitor will typically improve the 2nd harmonic distortion performance by
3dB to 6dB.
Figure 2 shows the AC-coupled, gain of +2, single-supply
circuit configuration used as the basis of the +5V Specifications and Typical Performance Curves. Though not a “railto-rail” design, the OPA3681 requires minimal input and
output voltage headroom compared to other very wideband
current-feedback op amps. It will deliver a 3Vp-p output
swing on a single +5V supply with greater than 150MHz
bandwidth. The key requirement of broadband single-supply
operation is to maintain input and output signal swings
within the usable voltage ranges at both the input and the
output. The circuit of Figure 2 establishes an input midpoint
bias using a simple resistive divider from the +5V supply
(two 806Ω resistors). The input signal is then AC-coupled
into this midpoint voltage bias. The input voltage can swing
to within 1.5V of either supply pin, giving a 2Vp-p input
signal range centered between the supply pins. The input
impedance matching resistor (57.6Ω) used for testing is
adjusted to give a 50Ω input match when the parallel
combination of the biasing divider network is included. The
gain resistor (RG) is AC-coupled, giving the circuit a DC
gain of +1, which puts the input DC bias voltage (2.5V) on
the output as well. Again, on a single +5V supply, the output
voltage can swing to within 1V of either supply pin while
delivering more than 75mA output current. A demanding
100Ω load to a midpoint bias is used in this characterization
circuit. The new output stage used in the OPA3681 can
deliver large bipolar output currents into this midpoint load
with minimal crossover distortion, as shown by the +5V
supply, 3rd harmonic distortion plots.
+5V
+VS
6.8µF
+
+
0.1µF
50Ω Source
DIS
VI
6.8µF
806Ω
50Ω
VO
1/3
OPA3681
0.1µF
50Ω
0.1µF
50Ω Load
VI
57.6Ω
DIS
806Ω
1/3
OPA3681
VO
100Ω
VS/2
RF
499Ω
RF
499Ω
RG
499Ω
RG
499Ω
+
6.8µF
0.1µF
0.1µF
–VS
–5V
FIGURE 1. DC-Coupled, G = +2, Bipolar Supply, Specification and Test Circuit.
OPA3681
SBOS095A
FIGURE 2. AC-Coupled, G = +2, Single Supply, Specification and Test Circuit.
11
TRIPLE ADC BUFFER CHANNEL
The OPAx681 family is ideally suited to single supply,
wideband ADC driving. A current feedback op amp is ideal
where high gains with high bandwidths are required. The
wide 3Vp-p output swing with over 150MHz full power
bandwidth on a single +5V supply is well suited to the
2Vp-p input range commonly required from modern CMOS
pipelined ADCs. Three channels of very high speed digitizer
channels are shown in Figure 3 using the OPA3681 driving
three ADS831s (8-bit, 80Msps CMOS converters). Each
input is AC-coupled into a 50Ω gain resistor that also will
act as a 50Ω impedance match at high frequencies. The
amplifier’s inputs and outputs are centered on the ADC
common-mode input voltage by tying each converter’s VCM
to the non-inverting inputs of the amplifier. This VCM acts as
the swing midpoint for the input to the converter. Since the
ADS831 can operate with differential inputs, driving into
the IN input will give a net non-inverting signal channel
even with the amplifiers operating at an inverting gain of
–6. The other input to the ADS831 is tied to this VCM as well
to give an input signal midpoint equal to VCM. The 300Ω
feedback resistor will be the output load in this configuration. Harmonic distortion for the OPA3681 will not degrade
the converter’s SFDR performance in this application.
WIDEBAND RGB MULTIPLEXER
The OPA3681 is ideally suited to implementing a simple,
very wideband, 2x1 RGB multiplexer. This simple “wiredOR video multiplexer” can be easily implemented using the
circuit shown in Figure 4.
This circuit uses two OPA3681s where each package accepts the three RGB component video signals from one of
two possible sources. Each non-inverting input is terminated
+5V
VDIS
+5V
Power Supply
De-Coupling Not Shown
U1
R1
75Ω
82.5Ω
1/3
OPA3681
VOUT Red
75Ω Line
340Ω
402Ω
G1
75Ω
82.5Ω
1/3
OPA3681
VOUT Green
75Ω Line
340Ω
Power Supply
De-Coupling
Not Shown
+5V
300Ω
IN
VCM
B1
75Ω
0.1µF
402Ω
1/3
OPA3681
22Ω
IN
47pF
ADS831
8-Bit
80Msps
82.5Ω
1/3
OPA3681
VOUT Blue
75Ω Line
340Ω
402Ω
–5V
0.1µF 50Ω
300Ω
+5V
V1
U2
300Ω
R2
IN
75Ω
VCM
0.1µF
1/3
OPA3681
22Ω
IN
47pF
ADS831
8-Bit
80Msps
1/3
OPA3681
340Ω
402Ω
G2
0.1µF 50Ω
75Ω
300Ω
82.5Ω
1/3
OPA3681
82.5Ω
V1
300Ω
0.1µF
1/3
OPA3681
22Ω
IN
47pF
340Ω
IN
VCM
ADS831
8-Bit
80Msps
402Ω
B2
75Ω
1/3
OPA3681
340Ω
0.1µF 50Ω
402Ω
300Ω
V1
FIGURE 3. ADC Driver.
12
82.5Ω
–5V
FIGURE 4. Wideband 2x1 RGB Multiplexer.
OPA3681
SBOS095A
in 75Ω to match the typical video source impedance. The
disable control is used to switch between channels by feeding a logic control line directly to all three VDIS inputs on
one package, and its complement to the three VDIS inputs on
the other. Since the disable feature is intentionally makebefore-break (to ensure that the output does not float in
transition), each of the two possible outputs for the three
RGB lines are combined through a limiting resistor. This
82.5Ω resistor limits the current between the two outputs
during switching. Each output will have a disabled channel.
The feedback and output network connected on the output
slightly attenuates the signal going out onto the 75Ω cable.
The gain and output matching resistors (82.5Ω) have been
slightly increased to get a signal gain of +1 to the matched
load and provide a 75Ω output impedance to the cable. The
section on Disable Operation shows the turn-on and turn-off
switching glitches, using a grounded input for the single
channel, is typically less than ±50mV. Where two outputs
are switched (shown in Figure 4), the output line is always
under the control of one amplifier or the other due to the
“make-before-break” disable timing. In this case, the switching glitches for 0V inputs drops to < 20mV.
The first op amp buffers the video DAC output and the first
filter section from each other. This first filter section provides group delay equalization. The second and third filter
sections provide a 6th-order lowpass filter response that also
compensates for the DAC’s sin(x)/x response.
VIDEO DAC RECONSTRUCTION FILTER
Wideband current-feedback op amps make ideal elements
for implementing high-speed active filters where the amplifier is used as fixed gain block inside a passive RC circuit
network. Their relatively constant bandwidth versus gain,
provides low interaction between the actual filter poles and
the required gain for the amplifier. Figure 5 shows an
example of a video DAC reconstruction filter.
FIGURE 6. DAC Reconstruction Filter Response.
The delay-equalized filter in Figure 5 compensates for the
DAC’s sin(x)/x response, and minimizes aliasing artifacts. It
is designed for single +5V operation, with a 13.5Msps DAC
sampling rate, and a 5.5MHz cutoff frequency.
The filter response can be seen in Figure 6.
20
f–3dB
10
0
(dB)
–10
–20
–30
–40
–50
0
1
10
100
Frequency (MHz)
HIGH POWER xDSL LINE DRIVER
Emerging broadband access technologies are making significant demands on the output stage drivers. Some of the
higher frequency versions, particularly in VDSL, require
passive bandpass filters to spectrally isolate the upstream
from downstream frequency bands. Figure 7 shows one
possible implementation of this using single-ended filters
and giving differential push/pull drive into a transformer.
The DAC output from the analog front end (AFE) typically
requires isolation from the complex filter impedance. The
first stage provides a tunable gain (using RG) with a fixed
100pF
Video
100µF 499Ω
In
100pF
+5V
499Ω
97.6Ω
237Ω
220pF
+5V
402Ω
+5V
56pF
1/3
OPA3681
82.5Ω
243Ω
220pF
120pF
1/3
OPA3681
412Ω
56pF
1/3
OPA3681
75.5Ω
VO
499Ω
499Ω
499Ω
953Ω
+5V
100µF
953Ω
FIGURE 5. Filter Schematic.
OPA3681
SBOS095A
13
termination for the DAC, RT. It is very useful from a
distortion standpoint to scale the characteristic impedance
up for the filter. This reduces the loading at the first stage
amplifier output, typically improving 3rd-order terms directly, as well as some improvement in 2nd-order terms.
Figure 7 assumes a 100Ω characteristic impedance for the
filter. The filter is driven from a 100Ω source resistor into a
100Ω load that is formed by the input gain resistor of the
inverting amplifier channel. The other non-inverting input is
isolated by a series 50Ω resistor—principally to isolate that
input from the out-of-band source impedance of the filter. In
this example, the output stage is set up for a differential gain
of 8. The total gain from the output of the bandpass filter to
the line will be 4 • n, where n is the turns ratio used in the
transformer. Very broad bandwidths at high power levels are
possible using the OPA3681 in the circuit of Figure 7.
Recognize also, that the output is in fact bandlimited by the
filter. Very high dynamic range is possible inside the filter
bandwidth due to the significant performance margin provided by the OPA3681.
WIDEBAND DIFFERENTIAL AMPLIFIER
The differential amplifier (three amplifier instrumentation
topology) on the front page of this data sheet shows a
common application applied to this triple current feedback
op amp. The two input stage amplifiers are configured for a
relatively high differential gain of 10. Lowering the feedback resistor values in this input stage provides > 120MHz
bandwidth, even at this high gain setting. The signal is
applied to the high impedance, non-inverting inputs at the
input stage. The differential gain is set by (1 + 2RF/RG) = 10
using the values shown on the front page. The third amplifier
performs the differential-to-single-ended conversion in a
standard single op amp differential stage. This differential
stage, built using the 3rd wideband current-feedback op
amp, in the OPA3681 will give lower CMRR at DC than
using a voltage feedback part, but higher CMRR at higher
frequencies. Measured performance, with no resistor value
tuning, gave approximately 75dB at DC and > 55dB CMRR
(input referred) through 10MHz. To maintain good distortion performance for the input stage amplifiers, the loading
at each output has been matched while achieving the gain of
1 and differential characteristic of the output stage. To
improve DC CMRR, tune the resistor to ground at the noninverting input of the output stage amplifier.
WIDEBAND PROGRAMMABLE GAIN
By tying all three inputs together from a single source, and
all three outputs together to drive a common load, a very
wideband, programmable gain function may be implemented.
Figure 8 shows an example of this application where the
three channels have been set up for gains of 2, 4, and 8 to
their output pins. When driving a doubly-terminated 50Ω
load, this gives a user-selectable gain of 1, 2 and 4 to the
matched load. The feedback resistor value has been optimized for maximum flat bandwidth in each channel. This
will give an almost constant > 200MHz bandwidth at any of
the three gain settings. The desired gain is selected by using
the disable control lines to choose one of the three possible
amplifiers as the active channel. An additional 10Ω resistor
was included inside the loop on each output stage to limit
output stage currents if more than one output is on during
gain select transition. This will reduce the maximum available output voltage swing into the 100Ω total load shown in
Figure 8 to approximately ±3.2V, but will provide surge
current protection during channel switching. The 20Ω series
resistors on each non-inverting input serves to isolate the
input parasitic capacitance from the source.
50Ω
1/3
OPA3681
+5V
Supply De-Coupling
Not Shown
DSL
AFE
RT
1/3
OPA3681
400Ω
–5V
400Ω
RS
1:n
100Ω
133Ω
Bandpass
Filter
100Ω
RS
400Ω
1/3
OPA3681
RG
FIGURE 7. Single-to-Differential xDSL Line Driver.
14
OPA3681
SBOS095A
+5V
74HC238
Power Supply
De-Coupling Not Shown
+5V
Y0
D1
Y1
D2
20Ω
Y2
G = +2
1/3
OPA3681
499Ω
499Ω
20Ω
50Ω Load
G = +4
VIN
1/3
OPA3681
50Ω
10Ω
140Ω
10Ω
50Ω
422Ω
20Ω
G = +8
1/3
OPA3681
35.7Ω
10Ω
249Ω
–5V
FIGURE 8. Wideband Programmable Gain.
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO
OPTIMIZE BANDWIDTH
A current-feedback op amp like the OPA3681 can hold an
almost constant bandwidth over signal gain settings with the
proper adjustment of the external resistor values. This is shown
in the Typical Performance Curves; the small signal bandwidth
decreases only slightly with increasing gain. These curves also
show that the feedback resistor has been changed for each gain
setting. The resistor “values” on the inverting side of the circuit
for a current feedback op amp can be treated as frequency
response compensation elements while their “ratios” set the
signal gain. Figure 9 shows the small-signal frequency response
analysis circuit for the OPA3681.
VI
α
VO
RI
iERR
Z(S) iERR
RF
RG
The key elements of this current-feedback op amp model are:
α → Buffer gain from the non-inverting input to the inverting input
RI → Buffer output impedance
FIGURE 9. Current Feedback Transfer Function Analysis
Circuit.
iERR → Feedback error current signal
Z(s) → Frequency dependent open loop transimpedance gain from iERR to VO
The buffer gain is typically very close to 1.00 and is
normally neglected from signal gain considerations. It will,
however, set the CMRR for a single op amp differential
OPA3681
SBOS095A
amplifier configuration. For a buffer gain α < 1.0, the
CMRR = –20 • log (1– α) dB.
RI, the buffer output impedance, is a critical portion of the
bandwidth control equation. The OPA3681 is typically 42Ω.
15
VO
=
VI
1+
Eq. 1
R
α 1 + F
RG
α NG
=
R F 1 + R F + R I NG
R F + R I 1 +
Z ( S)
RG
Z ( S)
RF
NG ≡ 1 +
R G
This is written in a loop gain analysis format where the
errors arising from a non-infinite open-loop gain are shown
in the denominator. If Z(s) were infinite over all frequencies,
the denominator of Equation 1 would reduce to 1 and the
ideal desired signal gain shown in the numerator would be
achieved. The fraction in the denominator of Equation 1
determines the frequency response. Equation 2 shows this as
the loop gain equation:
Eq. 2
Z ( S)
= Loop Gain
R F + R I NG
If 20 • log (RF + NG • RI) were drawn on top of the openloop transimpedance plot, the difference between the two
would be the loop gain at a given frequency. Eventually,
Z(s) rolls off to equal the denominator of Equation 2 at
which point the loop gain has reduced to 1 (and the curves
have intersected). This point of equality is where the
amplifier’s closed-loop frequency response, given by Equation 1, will start to roll off and is exactly analogous to the
frequency at which the noise gain equals the open-loop
voltage gain for a voltage-feedback op amp. The difference
here is that the total impedance in the denominator of
Equation 2 may be controlled somewhat separately from the
desired signal gain (or NG).
The OPA3681 is internally compensated to give a maximally flat frequency response for RF = 499Ω at NG = 2 on
±5V supplies. Evaluating the denominator of Equation 2
(which is the feedback transimpedance) gives an optimal
target of 589Ω. As the signal gain changes, the contribution
of the NG • RI term in the feedback transimpedance will
change, but the total can be held constant by adjusting RF.
Equation 3 gives an approximate equation for optimum RF
over signal gain:
Eq. 3
R F = 589Ω – NG R I
As the desired signal gain increases, this equation will
eventually predict a negative RF. A somewhat subjective
limit to this adjustment can also be set by holding RG to a
16
minimum value of 20Ω. Lower values will load both the
buffer stage at the input and the output stage if RF gets too
low—actually decreasing the bandwidth. Figure 10 shows
the recommended RF vs NG for both ±5V and a single +5V
operation. The values shown in Figure 10 give a good
starting point for design where bandwidth optimization is
desired.
FEEDBACK RESISTOR vs NOISE GAIN
600
500
Feedback Resistor (Ω)
A current-feedback op amp senses an error current in the
inverting node (as opposed to a differential input error
voltage for a voltage feedback op amp) and passes this on to
the output through an internal frequency dependent
transimpedance gain. The Typical Performance Curves show
this open-loop transimpedance response. This is analogous
to the open-loop voltage gain curve for a voltage-feedback
op amp. Developing the transfer function for the circuit of
Figure 9 gives Equation 1:
+5V
400
±5V
300
200
100
0
0
5
10
15
20
Noise Gain
FIGURE 10. Recommended Feedback Resistor vs Noise
Gain.
The total impedance going into the inverting input may be
used to adjust the closed-loop signal bandwidth. Inserting a
series resistor between the inverting input and the summing
junction will increase the feedback impedance (denominator
of Equation 2), decreasing the bandwidth. The internal
buffer output impedance for the OPA3681 is slightly influenced by the source impedance looking out of the noninverting input terminal. High source resistors will have the
effect of increasing RI, decreasing the bandwidth. For those
single-supply applications which develop a midpoint bias at
the non-inverting input through high valued resistors, the
decoupling capacitor is essential for power supply ripple
rejection, non-inverting input noise current shunting, and to
minimize the high frequency value for RI in Figure 9.
INVERTING AMPLIFIER OPERATION
Since the OPA3681 is a general purpose, wideband currentfeedback op amp, most of the familiar op amp application
circuits are available to the designer. Those triple op amp
applications that require considerable flexibility in the feedback element (e.g., integrators, transimpedance, some filters) should consider the unity gain stable voltage-feedback
OPA2680, since the feedback resistor is the compensation
element for a current feedback op amp. Wideband inverting
operation (especially summing) is particularly suited to the
OPA3681. Figure 11 shows a typical inverting configuration
where the I/O impedances and signal gain from Figure 1 are
retained in an inverting circuit configuration.
OPA3681
SBOS095A
+5V
Power supply
de-coupling
not shown
50Ω Load
DIS
1/3
OPA3681
50Ω
Source
VO
50Ω
RF
464Ω
RG
226Ω
VI
RM
64.9Ω
–5V
FIGURE 11. Inverting Gain of –2 with Impedance Matching.
In the inverting configuration, two key design considerations must be noted. The first is that the gain resistor (RG)
becomes part of the signal channel input impedance. If input
impedance matching is desired (which is beneficial whenever the signal is coupled through a cable, twisted pair, long
PC board trace or other transmission line conductor), it is
normally necessary to add an additional matching resistor to
ground. RG by itself is normally not set to the required input
impedance since its value, along with the desired gain, will
determine a RF which may be non-optimal from a frequency
response standpoint. The total input impedance for the
source becomes the parallel combination of RG and RM.
The second major consideration, touched on in the previous
paragraph, is that the signal source impedance becomes part
of the noise gain equation and will have slight effect on the
bandwidth through Equation 1. The values shown in Figure
11 have accounted for this by slightly decreasing RF (from
Figure 1) to re-optimize the bandwidth for the noise gain of
Figure 11 (NG = 2.82) In the example of Figure 11, the RM
value combines in parallel with the external 50Ω source
impedance, yielding an effective driving impedance of
50Ω || 64Ω = 28.1Ω. This impedance is added in series with
RG for calculating the noise gain—which gives NG = 2.82.
This value, along with the RF of Figure 10 and the inverting
input impedance of 45Ω, are inserted into Equation 3 to get
a feedback transimpedance nearly equal to the 589Ω optimum value.
Note that the non-inverting input in this bipolar supply
inverting application is connected directly to ground. It is
often suggested that an additional resistor be connected to
ground on the non-inverting input to achieve bias current
error cancellation at the output. The input bias currents for
a current feedback op amp are not generally matched in
either magnitude or polarity. Connecting a resistor to ground
on the non-inverting input of the OPA3681 in the circuit of
Figure 11 will actually provide additional gain for that
input’s bias and noise currents, but will not decrease the
output DC error since the input bias currents are not matched.
OPA3681
SBOS095A
OUTPUT CURRENT AND VOLTAGE
The OPA3681 provides output voltage and current capabilities that are unsurpassed in a low cost dual monolithic op
amp. Under no-load conditions at 25°C, the output voltage
typically swings closer than 1V to either supply rail; the
guaranteed swing limit is within 1.2V of either rail. Into a
15Ω load (the minimum tested load), it is guaranteed to
deliver more than ±135mA.
The specifications described above, though familiar in the
industry, consider voltage and current limits separately. In
many applications, it is the voltage • current, or V-I product,
which is more relevant to circuit operation. Refer to the
“Output Voltage and Current Limitations” plot in the Typical Performance Curves. The X and Y axes of this graph
show the zero-voltage output current limit and the zerocurrent output voltage limit, respectively. The four quadrants give a more detailed view of the OPA3681’s output
drive capabilities, noting that the graph is bounded by a
“Safe Operating Area” of 1W maximum internal power
dissipation. Superimposing resistor load lines onto the plot
shows that the OPA3681 can drive ±2.5V into 25Ω or ±3.5V
into 50Ω without exceeding the output capabilities or the
1W dissipation limit. A 100Ω load line (the standard test
circuit load) shows the full ±3.9V output swing capability,
as shown in the Specifications Table.
The minimum specified output voltage and current over
temperature are set by worst-case simulations at the cold
temperature extreme. Only at cold startup will the output
current and voltage decrease to the numbers shown in the
guaranteed tables. As the output transistors deliver power,
their junction temperatures will increase, decreasing their
VBE’s (increasing the available output voltage swing) and
increasing their current gains (increasing the available output current). In steady state operation, the available output
voltage and current will always be greater than that shown
in the over-temperature specifications since the output stage
junction temperatures will be higher than the minimum
specified operating ambient.
To maintain maximum output stage linearity, no output
short-circuit protection is provided. This will not normally
be a problem since most applications include a series matching resistor at the output that will limit the internal power
dissipation if the output side of this resistor is shorted to
ground. However, shorting the output pin directly to the
adjacent positive power supply pins will, in most cases,
destroy the amplifier. If additional short-circuit protection
is required, consider a small series resistor in the power
supply leads. Under heavy output loads, this will reduce the
available output voltage swing. A 5Ω series resistor in each
power supply lead will limit the internal power dissipation to
less than 1W for an output short circuit while decreasing the
available output voltage swing only 0.5V for up to 100mA
desired load currents. Always place the 0.1µF power supply
decoupling capacitors after these supply current-limiting
resistors directly on the supply pins.
17
DRIVING CAPACITIVE LOADS
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an A/D converter—including
additional external capacitance which may be recommended
to improve A/D linearity. A high speed, high open-loop gain
amplifier like the OPA3681 can be very susceptible to
decreased stability and closed-loop response peaking when
a capacitive load is placed directly on the output pin. When
the amplifier’s open-loop output resistance is considered,
this capacitive load introduces an additional pole in the
signal path that can decrease the phase margin. Several
external solutions to this problem have been suggested.
When the primary considerations are frequency response
flatness, pulse response fidelity and/or distortion, the simplest and most effective solution is to isolate the capacitive
load from the feedback loop by inserting a series isolation
resistor between the amplifier output and the capacitive
load. This does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher
frequency. The additional zero acts to cancel the phase lag
from the capacitive load pole, thus increasing the phase
margin and improving stability.
The Typical Performance Curves show the recommended
RS vs Capacitive Load and the resulting frequency response
at the load. Parasitic capacitive loads greater than 2pF can
begin to degrade the performance of the OPA3681. Long PC
board traces, unmatched cables, and connections to multiple
devices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA3681 output
pin (see Board Layout Guidelines).
DISTORTION PERFORMANCE
The OPA3681 provides good distortion performance into a
100Ω load on ±5V supplies. Relative to alternative solutions, it provides exceptional performance into lighter loads
and/or operating on a single +5V supply. Generally, until the
fundamental signal reaches very high frequency or power
levels, the 2nd harmonic will dominate the distortion with a
negligible 3rd harmonic component. Focusing then on the
2nd harmonic, increasing the load impedance improves
distortion directly. Remember that the total load includes
the feedback network; in the non-inverting configuration
(Figure 1), this is the sum of RF + RG, while in the inverting
configuration it is just RF. Also, providing an additional
supply decoupling capacitor (0.1µF) between the supply
pins (for bipolar operation) improves the 2nd-order distortion slightly (3dB to 6dB).
In most op amps, increasing the output voltage swing increases harmonic distortion directly. The Typical Performance
Curves show the 2nd harmonic increasing at a little less than
the expected 2x rate while the 3rd harmonic increases at a little
less than the expected 3x rate. Where the test power doubles,
the difference between it and the 2nd harmonic decreases less
than the expected 6dB while the difference between it and the
3rd decreases by less than the expected 12dB. This also shows
18
up in the 2-tone, 3rd-order intermodulation spurious (IM3)
response curves. The 3rd- order spurious levels are extremely
low at low output power levels. The output stage continues to
hold them low even as the fundamental power reaches very
high levels. As the Typical Performance Curves show, the
spurious intermodulation powers do not increase as predicted
by a traditional intercept model. As the fundamental power
level increases, the dynamic range does not decrease significantly. For 2 tones centered at 20MHz, with 10dBm/tone into
a matched 50Ω load (i.e., 2Vp-p for each tone at the load,
which requires 8Vp-p for the overall 2-tone envelope at the
output pin), the Typical Performance Curves show 62dBc
difference between the test tone power and the 3rd-order
intermodulation spurious levels. This exceptional performance improves further when operating at lower frequencies.
NOISE PERFORMANCE
Wideband current feedback op amps generally have a higher
output noise than comparable voltage-feedback op amps.
The OPA3681 offers an excellent balance between voltage
and current noise terms to achieve low output noise. The
inverting current noise (15pA/√Hz) is significantly lower
than earlier solutions while the input voltage noise
(2.2nV/√Hz) is lower than most unity gain stable, wideband,
voltage feedback op amps. This low input voltage noise was
achieved at the price of higher non-inverting input current
noise (12pA/√Hz). As long as the AC source impedance
looking out of the non-inverting node is less than 100Ω, this
current noise will not contribute significantly to the total
output noise. The op amp input voltage noise and the two
input current noise terms combine to give low output noise
under a wide variety of operating conditions. Figure 12
shows the op amp noise analysis model with all the noise
terms included. In this model, all noise terms are taken to be
noise voltage or current density terms in either nV/√Hz or
pA/√Hz.
ENI
1/3
OPA3681
RS
EO
IBN
ERS
RF
√ 4kTRS
4kT
RG
RG
IBI
√ 4kTRF
4kT = 1.6E –20J
at 290°K
FIGURE 12. Op Amp Noise Analysis Model.
OPA3681
SBOS095A
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 4 shows the general form for the
output noise voltage using the terms shown in Figure 12.
Eq. 4
EO =
(E
2
NI
)
+ (I BN R S ) + 4 kTR S NG 2 + (I BI R F ) + 4 kTR F NG
2
2
DISABLE OPERATION
The OPA3681 provides an optional disable feature that may
be used either to reduce system power or to implement a
simple channel multiplexing operation. If the DIS control
pin is left unconnected, the OPA3681 will operate normally.
To disable, the control pin must be asserted low. Figure 13
shows a simplified internal circuit for the disable control
feature.
+VS
Dividing this expression by the noise gain (NG = (1+RF /RG ))
will give the equivalent input referred spot noise voltage at the
non-inverting input as shown in Equation 5.
15kΩ
Eq. 5
E N = E NI + (I BN R S )
2
2
Q1
I R 2 4 kTR F
+ 4 kTR S + BI F +
NG
NG
Evaluating these two equations for the OPA3681 circuit and
component values shown in Figure 1 will give a total output
spot noise voltage of 8.4nV/√Hz and a total equivalent input
spot noise voltage of 4.2nV/√Hz. This total input-referred
spot noise voltage is higher than the 2.2nV/√Hz specification for the op amp voltage noise alone. This reflects the
noise added to the output by the inverting current noise times
the feedback resistor. If the feedback resistor is reduced in
high gain configurations (as suggested previously), the total
input-referred voltage noise given by Equation 5 will approach just the 2.2nV/√Hz of the op amp itself. For example,
going to a gain of +10 using RF = 182Ω will give a total
input referred noise of 2.4nV/√Hz.
DC ACCURACY AND OFFSET CONTROL
A current-feedback op amp like the OPA3681 provides
exceptional bandwidth in high gains, giving fast pulse settling but only moderate DC accuracy. The Specifications
Table shows an input offset voltage comparable to highspeed, voltage-feedback amplifiers. However, the two input
bias currents are somewhat higher and are unmatched.
Whereas bias current cancellation techniques are very effective with most voltage-feedback op amps, they do not
generally reduce the output DC offset for wideband currentfeedback op amps. Since the two input bias currents are
unrelated in both magnitude and polarity, matching the
source impedance looking out of each input to reduce their
error contribution to the output is ineffective. Evaluating the
configuration of Figure 1, using worst-case +25°C input
offset voltage and the two input bias currents, gives a worstcase output offset range equal to:
± (NG • VOS(MAX)) + (IBN • RS /2 • NG) ± (IBI • RF)
where NG = non-inverting signal gain
= ± (2 • 5.0mV) + (55µA • 25Ω • 2) ± (499Ω • 40µA)
= ±10mV + 2.75mV ± 20mV
25kΩ
VDIS
110kΩ
IS
Control
–VS
FIGURE 13. Simplified Disable Control Circuit.
In normal operation, base current to Q1 is provided through
the 110kΩ resistor while the emitter current through the
15kΩ resistor sets up a voltage drop that is inadequate to
turn on the two diodes in Q1’s emitter. As VDIS is pulled
low, additional current is pulled through the 15kΩ resistor
eventually turning on these two diodes (≈ 100µA). At this
point, any further current pulled out of VDIS goes through
those diodes holding the emitter-base voltage of Q1 at
approximately zero volts. This shuts off the collector current
out of Q1, turning the amplifier off. The supply current in
the disable mode is that only required to operate the circuit
of Figure 13. Additional circuitry ensures that turn-on time
occurs faster than turn-off time (make-before-break).
When disabled, the output and input nodes go to a high
impedance state. If the OPA3681 is operating in a gain of
+1, this will show a very high impedance (4pF || 1MΩ) at the
output and exceptional signal isolation. If operating at a
gain greater than +1, the total feedback network resistance
(RF + RG) will appear as the impedance looking back into
the output, but the circuit will still show very high forward
and reverse isolation. If configured as an inverting amplifier, the input and output will be connected through the
feedback network resistance (RF + RG) giving relatively
poor input to output isolation.
One key parameter in disable operation is the output glitch
when switching in and out of the disable mode. Figure 14
shows these glitches for the circuit of Figure 1 with the input
signal set to zero volts. The glitch waveform at the output
pin is plotted along with the DIS pin voltage.
= –27.25mV → +32.75mV
OPA3681
SBOS095A
19
Output Voltage (20mV/div)
40
20
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a high frequency
amplifier like the OPA3681 requires careful attention to
board layout parasitics and external component types. Recommendations that will optimize performance include:
Output Voltage
(0V Input)
0
–20
–40
4.8V
VDIS
0.2V
Time (20ns/div)
FIGURE 14. Disable/Enable Glitch.
The transition edge rate (dv/dt) of the DIS control line will
influence this glitch. For the plot of Figure 14, the edge rate
was reduced until no further reduction in glitch amplitude
was observed. This approximately 1V/ns maximum slew
rate may be achieved by adding a simple RC filter into the
VDIS pin from a higher speed logic line. If extremely fast
transition logic is used, a 2kΩ series resistor between the
logic gate and the VDIS input pin will provide adequate
bandlimiting using just the parasitic input capacitance on the
VDIS pin while still ensuring adequate logic level swing.
THERMAL ANALYSIS
Due to the high output power capability of the OPA3681,
heatsinking or forced airflow may be required under extreme
operating conditions. Maximum desired junction temperature will set the maximum allowed internal power dissipation as described below. In no case should the maximum
junction temperature be allowed to exceed 175°C. Operating
junction temperature (TJ) is given by TA + PD • θJA. The total
internal power dissipation (PD) is the sum of quiescent
power (PDQ) and additional power dissipation in the output
stage (PDL) to deliver load power. Quiescent power is simply
the specified no-load supply current times the total supply
voltage across the part. PDL will depend on the required
output signal and load but would, for a grounded resistive
load, be at a maximum when the output is fixed at a voltage
equal to 1/2 of either supply voltage (for equal bipolar
supplies). Under this condition, PDL = VS2/(4 • RL) where RL
includes feedback network loading.
Note that it is the power in the output stage and not into the
load that determines internal power dissipation.
As a worst-case example, compute the maximum TJ using an
OPA3681 SO-16 (in the circuit of Figure 1), operating at the
maximum specified ambient temperature of +85°C with all
three outputs driving a grounded 20Ω load to +2.5V:
PD = 10V • 19.2mA + 3 • [52/(4 • (20Ω || 998Ω))] = 1.15W
Maximum TJ = +85°C + (1.15 • 100°C/W) = 200°C
This absolute worst-case condition exceeds specified maximum junction temperature. Normally this extreme case will
not be encountered. Careful attention to internal power
dissipation is required and perhaps airflow considered under
extreme conditions.
20
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the output
and inverting input pins can cause instability: on the noninverting input, it can react with the source impedance to
cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be
opened in all of the ground and power planes around those
pins. Otherwise, ground and power planes should be unbroken elsewhere on the board.
b) Minimize the distance (< 0.25") from the power supply
pins to high frequency 0.1µF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power supply
connections (on pins 4 and 7) should always be decoupled
with these capacitors. An optional supply de-coupling capacitor across the two power supplies (for bipolar operation)
will improve 2nd harmonic distortion performance. Larger
(2.2µF to 6.8µF) decoupling capacitors, effective at lower
frequency, should also be used on the main supply pins.
These may be placed somewhat farther from the device and
may be shared among several devices in the same area of the
PC board.
c) Careful selection and placement of external components will preserve the high frequency performance of
the OPA3681. Resistors should be a very low reactance
type. Surface-mount resistors work best and allow a tighter
overall layout. Metal-film and carbon composition, axially
leaded resistors can also provide good high frequency performance. Again, keep their leads and PC board trace length
as short as possible. Never use wirewound type resistors in
a high frequency application. Since the output pin and
inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other
network components, such as non-inverting input termination resistors, should also be placed close to the package.
Where double-side component mounting is allowed, place
the feedback resistor directly under the package on the other
side of the board between the output and inverting input
pins. The frequency response is primarily determined by the
feedback resistor value as described previously. Increasing
its value will reduce the bandwidth, while decreasing it will
give a more peaked frequency response. The 499Ω feedback
resistor used in the typical performance specifications at a
gain of +2 on ±5V supplies is a good starting point for
design. Note that a 549Ω feedback resistor, rather than a
direct short, is recommended for the unity gain follower
application. A current feedback op amp requires a feedback
resistor even in the unity gain follower configuration to
control stability.
OPA3681
SBOS095A
d) Connections to other wideband devices on the board
may be made with short direct traces or through on-board
transmission lines. For short connections, consider the trace
and the input to the next device as a lumped capacitive load.
Relatively wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up around
them. Estimate the total capacitive load and set RS from the
plot of recommended RS vs Capacitive Load. Low parasitic
capacitive loads (< 5pF) may not need an RS since the
OPA3681 is nominally compensated to operate with a 2pF
parasitic load. If a long trace is required, and the 6dB signal
loss intrinsic to a doubly-terminated transmission line is
acceptable, implement a matched impedance transmission
line using microstrip or stripline techniques (consult an ECL
design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary on
board, and in fact, a higher impedance environment will
improve distortion as shown in the Distortion vs Load plots.
With a characteristic board trace impedance defined based
on board material and trace dimensions, a matching series
resistor into the trace from the output of the OPA3681 is
used as well as a terminating shunt resistor at the input of the
destination device. Remember also that the terminating
impedance will be the parallel combination of the shunt
resistor and the input impedance of the destination device:
this total effective impedance should be set to match the
trace impedance. The high output voltage and current capability of the OPA3681 allows multiple destination devices to
be handled as separate transmission lines, each with their
own series and shunt terminations. If the 6dB attenuation of
a doubly-terminated transmission line is unacceptable, a
long trace can be series-terminated at the source end only.
Treat the trace as a capacitive load in this case and set the
series resistor value as shown in the plot of RS vs Capacitive
Load. This will not preserve signal integrity as well as a
doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation
due to the voltage divider formed by the series output into
the terminating impedance.
e) Socketing a high speed part like the OPA3681 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it
almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the OPA3681
onto the board.
INPUT AND ESD PROTECTION
The OPA3681 is built using a very high speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins have limited ESD
protection using internal diodes to the power supplies as
shown in Figure 15.
currents are possible (e.g., in systems with ±15V supply
parts driving into the OPA3681), current limiting series
resistors should be added into the two inputs. Keep these
resistor values as low as possible since high values degrade
both noise performance and frequency response.
+V CC
External
Pin
Internal
Circuitry
–V CC
FIGURE 15. Internal ESD Protection.
DESIGN-IN TOOLS
APPLICATIONS SUPPORT
The Texas Instruments Applications Department is available
for design assistance at phone number 1-800-548-6132
(US/Canada only). The TI web site (www.ti.com) has the
latest data sheets and other design aids.
DEMONSTRATION BOARDS
A PC board will be available to assist in the initial evaluation
of circuit performance of the OPA3681. This is available as
an unpopulated PCB with descriptive documentation. See
the demonstration board literature for more information. The
summary information for this board is shown below:
PRODUCT
PACKAGE
DEMONSTRATION
BOARD
OPA3681E
OPA3681U
SSOP-16
SO-16
DEM-OPA368xE
DEM-OPA368xU
LITERATURE
REQUEST
NUMBER
MKT-354
MKT-364
Check the TI web site for availability of these boards.
SPICE MODELS
Computer simulation of circuit performance using SPICE is
often useful when analyzing the performance of analog
circuits and systems. This is particularly true for high speed
active devices, like the OPA3681, where parasitic capacitance and inductance can have a major effect on frequency
response.
SPICE models will be available through the TI web page or
on a disk (call our Applications Department). These models
do a good job of predicting small-signal AC and transient
performance under a wide variety of operating conditions.
They do not do as well in predicting the harmonic distortion
or differential gain and phase characteristics. These models
do not distinguish between the AC performance of different
package types.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
OPA3681
SBOS095A
21
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Following are URLs where you can obtain information on other Texas Instruments products and application solutions:
Products
Amplifiers
Data Converters
DSP
Clocks and Timers
Interface
Logic
Power Mgmt
Microcontrollers
RFID
RF/IF and ZigBee® Solutions
amplifier.ti.com
dataconverter.ti.com
dsp.ti.com
www.ti.com/clocks
interface.ti.com
logic.ti.com
power.ti.com
microcontroller.ti.com
www.ti-rfid.com
www.ti.com/lprf
Applications
Audio
Automotive
Broadband
Digital Control
Medical
Military
Optical Networking
Security
Telephony
Video & Imaging
Wireless
www.ti.com/audio
www.ti.com/automotive
www.ti.com/broadband
www.ti.com/digitalcontrol
www.ti.com/medical
www.ti.com/military
www.ti.com/opticalnetwork
www.ti.com/security
www.ti.com/telephony
www.ti.com/video
www.ti.com/wireless
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