ISL6520AIB

DATASHEET ISL6520A FN9016 Rev 6.00 Dec 10, 2009 Single Synchronous Buck Pulse-Width Modulation (PWM) Controller The ISL6520A makes simple work out of implementing a complete control and protection scheme for a DC/DC stepdown converter. Designed to drive N-Channel MOSFETs in a synchronous buck topology, the ISL6520A integrates the control, output adjustment, monitoring and protection functions into a single 8 Lead package. Features The ISL6520A provides simple, single feedback loop, voltage-mode control with fast transient response. The output voltage can be precisely regulated to as low as 0.8V, with a maximum tolerance of ±1.5% over-temperature and line voltage variations. A fixed frequency oscillator reduces design complexity, while balancing typical application cost and efficiency. • Drives N-Channel MOSFETs The error amplifier features a 15MHz gain-bandwidth product and 8V/ms slew rate which enables high converter bandwidth for fast transient performance. The resulting PWM duty cycles range from 0% to 100%. • Lossless, Programmable Overcurrent Protection - Uses Upper MOSFET’s rDS(ON) Protection from overcurrent conditions is provided by monitoring the rDS(ON) of the upper MOSFET to inhibit PWM operation appropriately. This approach simplifies the implementation and improves efficiency by eliminating the need for a current sense resistor. Pinouts ISL6520A (8 LD SOIC) TOP VIEW • 0.8V to VIN Output Range - 0.8V Internal Reference - ±1.5% Over Line Voltage and Temperature • Simple Single-Loop Control Design - Voltage-Mode PWM Control • Fast Transient Response - High-Bandwidth Error Amplifier - Full 0% to 100% Duty Cycle • Small Converter Size - 300kHz Fixed Frequency Oscillator - Internal Soft-Start - 8 Ld SOIC or 16 Ld 4mmx4mm QFN • QFN Package: - Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat No Leads - Package Outline - Near Chip Scale Package footprint, which improves PCB efficiency and has a thinner profile • Pb-Free (RoHS Compliant) 8 PHASE BOOT 1 7 COMP/SD UGATE 2 6 FB GND 3 5 VCC LGATE 4 NC NC PHASE NC ISL6520A (16 LD QFN) TOP VIEW 16 15 14 13 GND 3 10 NC NC 4 9 7 8 NC 11 COMP/OCSET VCC 2 NC UGATE LGATE 12 NC 6 • Power Supplies for Microprocessors - PCs - Embedded Controllers • Subsystem Power Supplies - PCI/AGP/GTL+ Buses - ACPI Power Control - SSTL-2 and DDR SDRAM Bus Termination Supply • DSP and Core Communications Processor Supplies 1 5 Applications • Cable Modems, Set Top Boxes, and DSL Modems BOOT FN9016 Rev 6.00 Dec 10, 2009 • Operates from +5V Input FB • Memory Supplies • Personal Computer Peripherals • Industrial Power Supplies • 5V-Input DC/DC Regulators • Low-Voltage Distributed Power Supplies Page 1 of 12 ISL6520A Ordering Information PART NUMBER (Note) PART MARKING TEMP. RANGE (°C) PACKAGE (Pb-free) PKG. DWG. # ISL6520ACBZ* 6520 ACBZ 0 to +70 8 Ld SOIC M8.15 ISL6520ACBZA* 6520 ACBZ 0 to +70 8 Ld SOIC M8.15 ISL6520AIBZ* 6520 AIBZ -40 to +85 8 Ld SOIC M8.15 ISL6520ACRZ* 65 20ACRZ 0 to +70 16 Ld 4x4mm QFN L16.4x4 ISL6520AIRZ* 65 20AIRZ -40 to +85 16 Ld 4x4mm QFN L16.4x4 ISL6520EVAL1 Evaluation Board *Add “-T” suffix for tape and reel. Please refer to TB347 for details on reel specifications. NOTE: These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. FN9016 Rev 6.00 Dec 10, 2009 Page 2 of 12 ISL6520A Block Diagram VCC POR AND SOFTSTART + - SAMPLE AND HOLD BOOT OC COMPARATOR UGATE + 0.8V PWM COMPARATOR ERROR AMP + - - + - INHIBIT PHASE GATE CONTROL PWM LOGIC VCC FB LGATE COMP/OCSET 20A OSCILLATOR FIXED 300kHz GND Typical Application VCC CBULK CDCPL VCC ROCSET 5 COMP/OCSET 7 1 ISL6520A 2 8 RF CI CF CHF DBOOT 6 FB 4 3 BOOT CBOOT UGATE LOUT PHASE LGATE +VO COUT GND ROFFSET RS FN9016 Rev 6.00 Dec 10, 2009 Page 3 of 12 ISL6520A Absolute Maximum Ratings Thermal Information Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +6.0V Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . +15.0V Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . 7.0V (DC) . . . . . . . . . . . . . . . . . . . . . . . . . . 8.0V ( I OUT  MAX  + ---------- , 2 whereI is the output inductor ripple current. For an equation for the ripple current see the section under component guidelines titled “Output Inductor Selection” on page 8. Soft-Start The POR function initiates the soft-start sequence after the overcurrent set point has been sampled. Soft-start clamps the error amplifier output (COMP pin) and reference input (non-inverting terminal of the error amp) to the internally generated soft-start voltage. Figure 2 shows a typical start-up interval where the COMP/OCSET pin has been released from a grounded (system shutdown) state. Initially, the COMP/OCSET is used to sample the overcurrent setpoint by disabling the error amplifier and drawing 20µA through ROCSET. Once the overcurrent level has been sampled, the soft-start function is initiated. The clamp on the error amplifier (COMP/OCSET pin) initially controls the converter’s output voltage during soft-start. The oscillator’s triangular waveform is compared to the ramping error amplifier voltage. This generates PHASE pulses of increasing width that charge the output capacitor(s). When the internally generated soft-start voltage exceeds the feedback (FB pin) voltage, the output voltage is in regulation. This method provides a rapid and FN9016 Rev 6.00 Dec 10, 2009 TIME (2ms/DIV.) FIGURE 2. SOFT-START INTERVAL Current Sinking The ISL6520A incorporates a MOSFET shoot-through protection method which allows a converter to sink current as well as source current. Care should be exercised when designing a converter with the ISL6520A when it is known that the converter may sink current. When the converter is sinking current, it is behaving as a boost converter that is regulating it’s input voltage. This means that the converter is boosting current into the VCC rail, which supplies the bias voltage to the ISL6520A. If there is nowhere for this current to go, such as to other distributed loads on the VCC rail, through a voltage limiting protection device, or other methods, the capacitance on the VCC bus will absorb the current. This situation will allow voltage level of the VCC rail to increase. If the voltage level of the rail is boosted to a level that exceeds the maximum voltage rating of the ISL6520A, then the IC will experience an irreversible failure and the converter will no longer be operational. Ensuring that there is a path for the current to follow other than the capacitance on the rail will prevent this failure mode. Application Guidelines Layout Considerations As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible, using ground plane construction or single point grounding. Page 6 of 12 ISL6520A pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). VIN ISL6520A Q1 PHASE CIN Q2 LGATE PWM COMPARATOR VOUT CO +VIN Q1 VOUT PHASE VCC +5V Q2 CO COMP/OCSET CVCC GND FIGURE 4. PRINTED CIRCUIT BOARD SMALL SIGNAL LAYOUT GUIDELINES Feedback Compensation Figure 5 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier (Error Amp) output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a FN9016 Rev 6.00 Dec 10, 2009 ZIN ERROR AMP REFERENCE DETAILED COMPENSATION COMPONENTS ZFB C2 C1 VOUT ZIN C3 R2 R3 R1 COMP FB + ISL6520A REFERENCE FIGURE 5. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN The modulator transfer function is the small-signal transfer function of VOUT/VE/A . This function is dominated by a DC Gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR . The DC Gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage VOSC . Modulator Break Frequency Equations LO LOAD ROCSET ISL6520A CO ESR (PARASITIC) + Figure 4 shows the circuit traces that require additional layout consideration. Use single point and ground plane construction for the circuits shown. Minimize any leakage current paths on the COMP/OCSET pin and locate the resistor, ROSCET close to the COMP/OCSET pin because the internal current source is only 20µA. Provide local VCC decoupling between VCC and GND pins. Locate the capacitor, CBOOT as close as practical to the BOOT and PHASE pins. All components used for feedback compensation should be located as close to the IC a practical. CBOOT PHASE ZFB Figure 3 shows the critical power components of the converter. To minimize the voltage overshoot, the interconnecting wires indicated by heavy lines should be part of a ground or power plane in a printed circuit board. The components shown in Figure 3 should be located as close together as possible. Please note that the capacitors CIN and CO may each represent numerous physical capacitors. Locate the ISL6520A within 3 inches of the MOSFETs, Q1 and Q2 . The circuit traces for the MOSFETs’ gate and source connections from the ISL6520A must be sized to handle up to 1A peak current. D1 DRIVER VOUT VE/A FIGURE 3. PRINTED CIRCUIT BOARD POWER AND GROUND PLANES OR ISLANDS BOOT LO + VOSC RETURN +5V VIN DRIVER OSC LO LOAD UGATE 1 F LC = ------------------------------------------2 x L O x C O 1 F ESR = -------------------------------------------2 x ESR x C O (EQ. 4) The compensation network consists of the error amplifier (internal to the ISL6520A) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The following equations relate the compensation network’s poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figure 7. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth. 2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC). 3. Place 2ND Zero at Filter’s Double Pole. Page 7 of 12 ISL6520A 4. Place 1ST Pole at the ESR Zero. These requirements are generally met with a mix of capacitors and careful layout. 5. Place 2ND Pole at Half the Switching Frequency. 6. Check Gain against Error Amplifier’s Open-Loop Gain. 7. Estimate Phase Margin - Repeat if Necessary. Compensation Break Frequency Equations 1 F Z1 = -----------------------------------2 x R 2 x C 1 1 F P1 = -------------------------------------------------------- C 1 x C 2 2 x R 2 x  ----------------------  C1 + C2  1 F Z2 = ------------------------------------------------------2 x  R 1 + R 3  x C 3 1 F P2 = -----------------------------------2 x R 3 x C 3 (EQ. 5) Figure 6 shows an asymptotic plot of the DC/DC converter’s gain vs frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 6. Using the previously mentioned guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the graph of Figure 6 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. FZ1 FZ2 FP1 FP2 80 OPEN LOOP ERROR AMP GAIN GAIN (dB) 60 20 20LOG (R2/R1) 0 20LOG (VIN/DVOSC) MODULATOR GAIN -20 COMPENSATION GAIN -40 -60 High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the Equivalent Series Inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. Output Inductor Selection 100 40 Modern components and loads are capable of producing transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (Effective Series Resistance) and voltage rating requirements rather than actual capacitance requirements. CLOSED LOOP GAIN FLC 10 100 1K FESR 10K 100K 1M 10M FREQUENCY (Hz) FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN Component Selection Guidelines Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. FN9016 Rev 6.00 Dec 10, 2009 The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: I = VIN - VOUT Fs x L x VOUT VIN VOUT = I x ESR (EQ. 6) Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL6520A will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. Page 8 of 12 ISL6520A The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: tRISE = L x ITRAN VIN - VOUT tFALL = L x ITRAN VOUT (EQ. 7) where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. The worst case response time can be either at the application or removal of load. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2 . The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. For a through hole design, several electrolytic capacitors may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge currentrating. These capacitors must be capable of handling the surge-current at power-up. Some capacitor series available from reputable manufacturers are surge current tested. losses. The lower switch realizes most of the switching losses when the converter is sinking current (see the following equations ). These equations assume linear voltagecurrent transitions and do not adequately model power loss due the reverse-recovery of the upper and lower MOSFET’s body diode. The gate-charge losses are dissipated by the ISL6520A and don't heat the MOSFETs. However, large gatecharge increases the switching interval, tSW which increases the MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. Losses while Sourcing Current 2 1 P UPPER = Io  r DS  ON   D + ---  Io  V IN  t SW  F S 2 PLOWER = Io2 x rDS(ON) x (1 - D) Losses while Sinking Current PUPPER = Io2 x rDS(ON) x D 2 1 P LOWER = Io  r DS  ON    1 – D  + ---  Io  V IN  t SW  F S 2 Where: D is the duty cycle = VOUT / VIN , tSW is the combined switch ON and OFF time, and (EQ. 8) FS is the switching frequency. Given the reduced available gate bias voltage (5V), logic-level or sub-logic-level transistors should be used for both N-MOSFETs. Caution should be exercised with devices exhibiting very low VGS(ON) characteristics. The shootthrough protection present aboard the ISL6520A may be circumvented by these MOSFETs if they have large parasitic impedences and/or capacitances that would inhibit the gate of the MOSFET from being discharged below it’s threshold level before the complementary MOSFET is turned on. +5V VCC BOOT CBOOT UGATE The ISL6520A requires 2 N-Channel power MOSFETs. These should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. FN9016 Rev 6.00 Dec 10, 2009 +5V + VD - ISL6520A MOSFET Selection/Considerations In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses are the largest component of power dissipation for both the upper and the lower MOSFETs. These losses are distributed between the two MOSFETs according to duty factor. The switching losses seen when sourcing current will be different from the switching losses seen when sinking current. When sourcing current, the upper MOSFET realizes most of the switching DBOOT Q1 PHASE - + LGATE NOTE: VG-S  VCC -VD Q2 NOTE: VG-S  VCC GND FIGURE 7. UPPER GATE DRIVE BOOTSTRAP Figure 7 shows the upper gate drive (BOOT pin) supplied by a bootstrap circuit from VCC . The boot capacitor, CBOOT, develops a floating supply voltage referenced to the PHASE pin. The supply is refreshed to a voltage of VCC less the boot diode drop (VD) each time the lower MOSFET, Q2 , turns on. Page 9 of 12 ISL6520A ISL6520A DC/DC Converter Application Circuit Figure 8 shows an application circuit of a DC/DC Converter. Detailed information on the circuit, including a complete Bill-of-Materials and circuit board description, can be found in Application Note AN9932. +5V + CIN 2 x 330µF 0.1µF 2 x 1µF VCC ISL6520A 6.19k 5 D1 MONITOR AND PROTECTION 1 2 UGATE COMP/OCSET 7 REF 10.0k 8 PHASE 0.1µF Q1 L1 + - 470pF 8200pF OSC U1 4 + - FB 6 1.00k BOOT VOUT LGATE Q2 3 + COUT 3 x 330µF 0.1µF GND 3.16k 60.4 18000pF Component Selection Notes: CIN - Each 330µF 6.3WVDC, Sanyo 6TPB330M or Equivalent. COUT - Each 330µF 6.3WVDC, Sanyo 6TPB330M or Equivalent. D1 - 30mA Schottky Diode, MA732 or Equivalent L1 - 3.1H Inductor, Panasonic P/N ETQ-P6F2ROLFA or Equivalent. Q1 , Q2 - Intersil MOSFET; HUF76143. FIGURE 8. 5V to 3.3V 15A DC/DC CONVERTER © Copyright Intersil Americas LLC 2001-2009. All Rights Reserved. All trademarks and registered trademarks are the property of their respective owners. For additional products, see www.intersil.com/en/products.html Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted in the quality certifications found at www.intersil.com/en/support/qualandreliability.html Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com FN9016 Rev 6.00 Dec 10, 2009 Page 10 of 12 ISL6520A Package Outline Drawing L16.4x4 16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 6, 02/08 4X 1.95 4.00 12X 0.65 A B 13 6 PIN 1 INDEX AREA 6 PIN #1 INDEX AREA 16 1 4.00 12 2 . 10 ± 0 . 15 9 4 0.15 (4X) 5 8 TOP VIEW 0.10 M C A B +0.15 16X 0 . 60 -0.10 4 0.28 +0.07 / -0.05 BOTTOM VIEW SEE DETAIL "X" 0.10 C 1.00 MAX ( 3 . 6 TYP ) ( 2 . 10 ) C BASE PLANE SEATING PLANE 0.08 C SIDE VIEW ( 12X 0 . 65 ) ( 16X 0 . 28 ) C 0 . 2 REF 5 ( 16 X 0 . 8 ) 0 . 00 MIN. 0 . 05 MAX. DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. FN9016 Rev 6.00 Dec 10, 2009 Page 11 of 12 ISL6520A Small Outline Plastic Packages (SOIC) M8.15 (JEDEC MS-012-AA ISSUE C) N 8 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE INDEX AREA H 0.25(0.010) M B M INCHES E SYMBOL -B1 2 3 L SEATING PLANE -A- A D h x 45° -C- e A1 B 0.25(0.010) M C 0.10(0.004) C A M MIN MAX MIN MAX NOTES A 0.0532 0.0688 1.35 1.75 - A1 0.0040 0.0098 0.10 0.25 - B 0.013 0.020 0.33 0.51 9 C 0.0075 0.0098 0.19 0.25 - D 0.1890 0.1968 4.80 5.00 3 E 0.1497 0.1574 3.80 4.00 4 e  B S NOTES: 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. MILLIMETERS 0.050 BSC 1.27 BSC - H 0.2284 0.2440 5.80 6.20 - h 0.0099 0.0196 0.25 0.50 5 L 0.016 0.050 0.40 1.27 6 N  8 0° 8 8° 0° 7 8° Rev. 1 6/05 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. FN9016 Rev 6.00 Dec 10, 2009 Page 12 of 12