RT6252A/B
17V Input, 2A, ACOT® Buck Converter with Both FETs OC
Protection
General Description
Features
The RT6252A/B is a simple, easy-to-use, 2A
synchronous step-down DC-DC converter with an input
supply voltage range of 4.5V to 17V. The device
possesses an accurate reference voltage and
integrates low RDS(ON) power MOSFETs to achieve
high efficiency.
The RT6252A/B adopts Advanced Constant On-Time
(ACOT® ) control architecture to provide an ultrafast
transient response with few external components and
to operate in nearly constant switching frequency over
the line, load, and output voltage range. The RT6252A
operates in automatic PSM that maintains high
efficiency during light load operation. The RT6252B
operates in Forced PWM that helps meet tight voltage
regulation accuracy requirements.
The RT6252A/B senses both FETs current for a robust
over-current protection (OCP). The device features
cycle-by-cycle current limit protection to prevent the
device from the catastrophic damage in output short
circuit, over-current or inductor saturation conditions. A
built-in soft-start function prevents inrush current during
start-up. The device also includes input under-voltage
2A Converter Integrated 140m and 84m FETs
Input Supply Voltage Range : 4.5V to 17V
Output Voltage Range : 0.765V to 7V
Advanced Constant On-Time (ACOT ® ) Control
Ultrafast Transient Response
Optimized
for Low-ESR Ceramic Output
Capacitors
High Accuracy Feedback Reference Voltage : Typ.
1%
Optional for Operation Modes :
Automatically
(RT6252A)
Power
Saving
Forced PWM Mode (RT6252B)
Mode
(PSM)
Fixed Switching Frequency : 580kHz
Enable Control and Internally Fixed Soft-Start
Safe Start-Up from Pre-biased Output
Input Under-Voltage Lockout (UVLO)
Output Under-Voltage Protection (UVP) with
Hiccup Mode
High- /Low-side MOSFET OCP and OTP Function
RoHS Compliant and Halogen Free
lockout, output under-voltage protection, and
over-temperature protection (OTP) to provide safe and
smooth operation in all operating conditions.
Simplified Application Circuit
RT6252A/B
VIN
VIN
RBOOT
(Optional)
BOOT
CBOOT
CIN
L
VOUT
SW
Enable
EN
RFB1
GND
1
August 2021
COUT
FB
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RT6252A/B
Ordering Information
Applications
RT6252A/B
Package Type
J6F : TSOT-23-6 (FC)
H6F : SOT-563 (FC)
Lead Plating System
G : Green (Halogen Free and Pb Free)
UVP Option
H : Hiccup
PWM Operation Mode
A : Automatic PSM
B : Forced PWM
Set-Top Boxes
LCD TVs
Home Networking Devices
Surveillance
General Purpose
Pin Configuration
(TOP VIEW)
BOOT EN
6
FB
5
4
2
3
Note :
Richtek products are :
GND SW VIN
RoHS compliant and compatible with the current
requirements of IPC/JEDEC J-STD-020.
TSOT-23-6 (FC)
Suitable for use in SnPb or Pb-free soldering processes.
FB
EN
BOOT
6
5
4
1
2
3
VIN
SW
GND
Marking Information
RT6252AHGJ6F
34=DNN
34= : Product Code
DNN : Date Code
RT6252BHGJ6F
33=DNN
33= : Product Code
DNN : Date Code
SOT-563 (FC)
RT6252AHGH6F
01W
01 : Product Code
W : Date Code
RT6252BHGH6F
00W
00 : Product Code
W : Date Code
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RT6252A/B
Functional Pin Description
Pin No.
TSOT23-6(FC) SOT-563 (FC)
Pin Name
Pin Function
1
3
GND
Power ground.
2
2
SW
Switch node between the internal switch and the synchronous
rectifier. Connect this pin to the inductor and bootstrap capacitor.
3
1
VIN
Power input. The input voltage range is from 4.5V to 17V. Connect
input bypass capacitors directly to this pin and GND pins. The
MLCC with capacitance higher than 20F is recommended.
4
6
FB
Feedback voltage input. Connect this pin to the midpoint of the
external feedback resistive divider to set the output voltage of the
converter to the desired regulation level. The device regulates the
FB voltage at feedback reference voltage, typically 0.765V.
5
5
EN
Enable control input. Connect this pin to logic high enables the
device and connect this pin to GND disables the device.
6
4
BOOT
Bootstrap capacitor connection node to supply the high-side gate
driver. Connect a 0.1F ceramic capacitor between this pin and
the SW pin.
Functional Block Diagram
VIN
SW
+
EN
UVLO
-
VEN_TH
Internal
Regulator
VCC
PVCC
OnTime
REN_DN
BOOT
OC
UV
Protection
65%
Control
Soft-Start
Ramp
Gen.
+
+
-
FB
Gate Driver &
Dead-Time
Control
SW
Comparator
MIN OFF
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GND
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RT6252A/B
Operation
The RT6252A/B is a high-efficiency, synchronous
step-down DC-DC converter that can deliver up to 2A
output current from a 4.5V to 17V input supply.
Advanced Constant On-Time Control and PWM
Operation
The RT6252A/B adopts ACOT® control for its ultrafast
transient response, low external component counts and
stable with low ESR MLCC output capacitors. When
the feedback voltage falls below the feedback
reference voltage, the minimum off-time one-shot
(200ns, typ.) has timed out and the inductor current is
below the current limit threshold, then the internal
on-time one-shot circuitry is triggered and the high-side
switch is turn-on. Since the minimum off-time is short,
the device exhibits ultrafast transient response and
enables the use of smaller output capacitance.
The on-time is inversely proportional to input voltage
and directly proportional to output voltage to achieve
pseudo-fixed frequency over the input voltage range.
After the on-time one-shot timer expired, the high-side
switch is turned off and the low-side switch is turned on
until the on-time one-shot is triggered again. To
achieve stable operation with low-ESR ceramic output
capacitors, an internal ramp signal is added to the
feedback reference voltage to simulate the output
voltage ripple.
Power Saving Mode (RT6252A Only)
The RT6252A automatically enters power saving mode
(PSM) at light load to maintain high efficiency. As the
load current decreases, the inductor current ripple
valley eventually touches the zero current, which is the
boundary between continuous conduction and
discontinuous conduction modes. The low-side switch
is turned off when the zero inductor current is detected.
In this case, the output capacitor is only discharged by
load current so that the switching frequency decreases.
As the result, the light-load efficiency can be enhanced
due to lower switching loss.
Enable Control
The RT6252A/B provides an EN pin, as an external
chip enable control, to enable or disable the device. If
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VEN is held below a logic-low threshold voltage (VEN_L)
of the enable input (EN), the converter will disable
output voltage, that is, the converter is disabled and
switching is inhibited even if the VIN voltage is above
VIN under-voltage lockout threshold (VUVLO). During
shutdown mode, the supply current can be reduced to
ISHDN (10A or below). If the EN voltage rises above
the logic-high threshold voltage (VEN_H) while the VIN
voltage is higher than UVLO threshold, the device will
be turned on, that is, switching being enabled and
soft-start sequence being initiated. An internal resistor
REN_DN from EN to GND allows EN float to shutdown
the chip.
Soft-Start (SS)
The RT6252A/B provides an internal soft-start feature
for inrush control, and the output voltage starts to rise
in 0.3ms from EN rising edge. At power up, the internal
capacitor is charged by an internal current source to
generate a soft-start ramp voltage as a reference
voltage to the PWM comparator. The device will initiate
switching and the output voltage will smoothly ramp up
to its targeted regulation voltage only after this ramp
voltage is greater than the feedback voltage VFB to
ensure the converter has a smooth start-up from
pre-biased output.
VIN = 12V
VIN
VCC = 5V
VCC
EN
0.3ms
tSS
VOUT
Input Under-Voltage Lockout
In addition to the EN pin, the RT6252A/B also provides
enable control through the VIN pin. It features an
under-voltage lockout (UVLO) function that monitors
the internal linear regulator (VCC). If VEN rises above
VEN_H first, switching will still be inhibited until the VIN
voltage rises above VUVLO. It is to ensure that the
internal regulator is ready so that operation with
not-fully-enhanced internal MOSFET switches can be
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RT6252A/B
prevented. After the device is powered up, if the input
voltage VIN goes below the UVLO falling threshold
voltage (VUVLO VUVLO), this switching will be
inhibited; if VIN rises above the UVLO rising threshold
(VUVLO), the device will resume normal operation with a
complete soft-start.
Output Under-Voltage Protection and Hiccup Mode
The RT6252A/B includes output under-voltage
protection (UVP) against over-load or short-circuited
condition by constantly monitoring the feedback
voltage VFB. If VFB drops below the under-voltage
protection trip threshold (typically 65% of the internal
feedback reference voltage), the UV comparator will go
high to turn off both the internal high-side and low-side
MOSFET switches.
If the output under-voltage condition continues for a
period of time, the RT6252A/B will enter output
under-voltage protection with hiccup mode. During
hiccup mode, the IC will shut down for tHICCUP_OFF
(15ms), and then attempt to recover automatically for
tHICCUP_ON (1.8ms). Upon completion of the soft-start
sequence, if the fault condition is removed, the
converter will resume normal operation; otherwise,
such cycle for auto-recovery will be repeated until the
fault condition is cleared. The hiccup mode allows the
circuit to operate safely with low input current and
power dissipation, and then the converter resumes
normal operation as soon as the over-load or
short-circuit condition is removed.
MOSFETs and prevents the device from the
catastrophic
damage
in
output
short-circuit,
over-current or inductor saturation conditions.
The high-side MOSFET over-current protection is
achieved by an internal current comparator that
monitors the current in the high-side MOSFET during
each on-time. The switch current is compared with the
high-side switch peak-current limit (ILIM_H) after a
certain amount of delay when the high-side switch
being turned on each cycle. If an over-current condition
occurs, the converter will immediately turns off the
high-side switch and turns on the low-side switch to
prevent the inductor current exceeding the high-side
current limit.
The low-side MOSFET over-current protection is
achieved by measuring the inductor current through the
synchronous rectifier (low-side switch) during the
low-side on-time. Once the current rises above the
low-side switch valley current limit (ILIM_L), the on-time
one-shot will be inhibited until the inductor current
ramps down to the current limit level (ILIM_L), that is,
another on-time can only be triggered when the
inductor current goes below the low-side current limit. If
the output load current exceeds the available inductor
current (clamped by the low-side current limit), the
output capacitor needs to supply the extra current such
that the output voltage will begin to drop. If it drops
below the output under-voltage protection trip threshold,
the IC will stop switching to avoid excessive heat.
Negative Over-Current Limit
Output short
VOUT, 2V/Div
Fault condition removed
Resume normal operation
ISW , 4A/Div
VSW , 10/Div
10ms/Div
The Over-Current Protection
The RT6252A/B features cycle-by-cycle current-limit
protection on both the high-side and low-side
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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August 2021
The RT6252B is the part which is forced to PWM and
allows negative current operation.
In case of PWM operation, high negative current may
be generated as an external power source is tied to
output terminal unexpectedly. As the risk described
above, the internal circuit monitors negative current in
each on-time interval of low-side MOSFET and
compares it with NOC threshold.
Once the negative current exceeds the NOC threshold,
the low-side MOSFET is turned off immediately, and
then the high-side MOSFET will be turned on to
discharge the energy of output inductor. This behavior
can keep the valley of negative current at NOC
threshold to protect low-side MOSFET. However, the
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RT6252A/B
negative current can’t be limited at NOC threshold
anymore since minimum off-time is reached.
Thermal Shutdown
The RT6252A/B includes an over-temperature
protection (OTP) circuitry to prevent overheating due to
excessive power dissipation. The OTP will shut down
switching operation when junction temperature
exceeds a thermal shutdown threshold (TSD). Once the
junction temperature cools down by a thermal
shutdown hysteresis (TSD), the IC will resume normal
operation with a complete soft-start.
Note that the over-temperature protection is intended to
protect the device during momentary overload
conditions. The protection is activated outside of the
absolute maximum range of operation as a secondary
fail-safe and therefore should not be relied upon
operationally. Continuous operation above the
specified absolute maximum operating junction
temperature may impair the reliability of the device or
permanently damage the device.
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RT6252A/B
Absolute Maximum Ratings
(Note 1)
Supply Input Voltage, VIN ------------------------------------------------------------------------------------------ 0.3V to 20V
Enable Voltage, EN -------------------------------------------------------------------------------------------------- 0.3V to 20V
Switch Voltage, SW -------------------------------------------------------------------------------------------------- 0.3V to 20.3V
< 100ns ----------------------------------------------------------------------------------------------------------------- 5V to 25V
BOOT Voltage , BOOT---------------------------------------------------------------------------------------------- – 0.3V to 26V
BOOT to SW, VBOOT VSW --------------------------------------------------------------------------------------- 0.3V to 6V
Feedback Voltage, FB ---------------------------------------------------------------------------------------------- 0.3V to 6V
Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------------- 260C
Junction Temperature ----------------------------------------------------------------------------------------------- 150C
Storage Temperature Range -------------------------------------------------------------------------------------- 65C to 150C
ESD Ratings
(Note 2)
ESD Susceptibility
HBM (Human Body Model) ---------------------------------------------------------------------------------------- 2kV
Recommended Operating Conditions
(Note 3)
Supply Input Voltage ------------------------------------------------------------------------------------------------ 4.5V to 17V
Junction Temperature Range ------------------------------------------------------------------------------------- 40C to 125C
Thermal Information
(Note 4 and Note 5)
Thermal Parameter
TSOT-23-6 (FC)
SOT-563 (FC)
Unit
JA
Junction-to-ambient thermal resistance
(JEDEC standard)
88.7
104.3
C/W
JC(Top)
Junction-to-case (top) thermal resistance
76.9
62.1
C/W
JC(Bottom)
Junction-to-case (bottom) thermal resistance
6
8.4
C/W
JA(EVB)
Junction-to-ambient thermal resistance
(specific EVB)
63.3
82.3
C/W
JC(Top)
Junction-to-top characterization parameter
15.3
7.5
C/W
JB
Junction-to-board characterization parameter
34.66
47.38
C/W
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RT6252A/B
Electrical Characteristics
TSOT-23-6 (FC)
(VIN = 12V, TA = 25C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Voltage
VIN Supply Input Operating
Voltage
VIN
4.5
--
17
V
Under-Voltage Lockout
Threshold
VUVLO
3.7
4
4.3
V
Under-Voltage Lockout
Threshold Hysteresis
VUVLO
--
400
--
mV
Shutdown Current
ISHDN
VEN = 0V
--
--
4
µA
Quiescent Current
IQ
VEN = 2V, VFB = 0.8V
--
280
--
µA
--
1
--
ms
Soft-Start
Soft-Start Time
tSS
Enable Voltage
Enable Voltage Threshold
EN Pin Pull-Down
Resistance
VEN_H
EN high-level input voltage
1.16
1.25
1.34
VEN_L
EN low-level input voltage
1.01
1.1
1.19
REN_DN
EN pin resistance to GND, VEN = 12V
225
450
900
k
V
Feedback Voltage and Discharge Resistance
Feedback Threshold Voltage VFB
VOUT = 1.05V
758
765
772
mV
Feedback Input Current
IFB
VFB = 0.8V, TA = 25°C
0.1
0
0.1
A
High-Side On-Resistance
RDS(ON)_H
VBOOT – VSW = 4.8V
--
140
--
Low-Side On-Resistance
RDS(ON)_L
--
84
--
High-Side Switch Current
Limit
ILIM_H
--
5
--
Low-Side Switch Valley
Current Limit
ILIM_L
Internal MOSFET
mΩ
Current Limit
A
2.2
3.2
4.2
--
580
--
kHz
--
60
--
ns
VFB = 0.5V
--
200
260
ns
Hiccup detect
--
65
--
%
Switching Frequency
Switching Frequency
f SW
VOUT = 1.05V, PWM mode
On-Time Timer Control
Minimum On-Time
tON_MIN
Minimum Off-Time
tOFF_MIN
Output Under-Voltage Protections
UVP Trip Threshold
VUVP
Hiccup Power On-Time
tHICCUP_ON
--
1.8
--
Hiccup Power Off-Time
tHICCUP_OFF
--
15
--
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RT6252A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Thermal Shutdown
Thermal Shutdown
Threshold
TSD
--
155
--
Thermal Shutdown
Hysteresis
TSD
--
35
--
°C
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RT6252A/B
SOT-563 (FC)
(VIN = 12V, TA = 25C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Voltage
VIN Supply Input Operating
Voltage
VIN
4.5
--
17
V
Under-Voltage Lockout
Threshold
VUVLO
3.9
4.2
4.5
V
Under-Voltage Lockout
Threshold Hysteresis
VUVLO
--
420
--
mV
Shutdown Current
ISHDN
VEN = 0V
--
--
4
µA
Quiescent Current
IQ
VEN = 2V, VFB = 0.85V
--
295
--
µA
--
0.95
--
ms
Soft-Start
Soft-Start Time
tSS
Enable Voltage
Enable Voltage Threshold
EN Pin Pull-Down
Resistance
VEN_H
EN high-level input voltage
1.24
1.31
1.38
VEN_L
EN low-level input voltage
1.09
1.16
1.23
REN_DN
EN pin resistance to GND, VEN = 12V
225
450
900
k
V
Feedback Voltage and Discharge Resistance
Feedback Threshold Voltage VFB
VOUT = 1.05V
799
807
815
mV
Feedback Input Current
IFB
VFB = 0.85V, TA = 25°C
0.1
0
0.1
A
High-Side On-Resistance
RDS(ON)_H
VBOOT – VSW = 4.8V
--
140
--
Low-Side On-Resistance
RDS(ON)_L
--
84
--
High-Side Switch Current
Limit
ILIM_H
--
5
--
Low-Side Switch Valley
Current Limit
ILIM_L
2.5
3.35
4.2
--
580
--
kHz
--
60
--
ns
VFB = 0.5V
--
190
250
ns
Hiccup detect
--
65
--
%
Internal MOSFET
mΩ
Current Limit
A
Switching Frequency
Switching Frequency
f SW
VOUT = 1.05V, PWM mode
On-Time Timer Control
Minimum On-Time
tON_MIN
Minimum Off-Time
tOFF_MIN
Output Under-Voltage Protections
UVP Trip Threshold
VUVP
Hiccup Power On-Time
tHICCUP_ON
--
1.8
--
Hiccup Power Off-Time
tHICCUP_OFF
--
15
--
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RT6252A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Thermal Shutdown
Thermal Shutdown
Threshold
TSD
--
155
--
Thermal Shutdown
Hysteresis
TSD
--
35
--
°C
Note 1. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These
are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in
the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect
device reliability.
Note 2. Devices are ESD sensitive. Handling precaution is recommended.
Note 3. The device is not guaranteed to function outside its operating conditions.
Note 4. θJA and θJC are measured or simulated at TA = 25C based on the JEDEC 51-7 standard.
Note 5. θJA(EVB), ΨJC(TOP) and ΨJB are measured on a high effective-thermal-conductivity four-layer test board which is in size of
70mm x 50mm; furthermore, all layers with 1 oz. Cu. Thermal resistance/parameter values may vary depending on the
PCB material, layout, and test environmental conditions.
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RT6252A/B
Typical Application Circuit
RT6252A/B
VIN
CIN
10μF x 2
CIN
0.1μF
Enable
VIN
BOOT
RBOOT
(Optional)
CBOOT
0.1μF
L
VOUT
SW
EN
RFB1
GND
COUT
CFF
FB
RFB2
Table 1. Recommended Components Selection
VOUT (V)
RFB1 (k)
RFB2 (k)
CFF (pF)
L (H)
COUT (F)
52.3
10
10 to 100
3.3 to 4.7
20 to 68
33.2
30.9
10
10 to 100
3.3 to 4.7
20 to 68
2.5
22.6
21
10
10 to 100
3.3 to 4.7
20 to 68
1.8
13.7
12.4
10
10 to 100
2.2 to 4.7
20 to 68
1.5
9.53
8.66
10
--
2.2 to 4.7
20 to 68
1.2
5.76
4.87
10
--
2.2 to 4.7
20 to 68
1.0
3.09
2.4
10
--
2.2 to 4.7
20 to 68
TSOT-23-6 (FC)
SOT-563 (FC)
5.0
54.9
3.3
Note :
(1) Please do not use a CFF higher than 100pF due to the noise coupling consideration.
(2) Considering effective capacitance de-rating which is related to biased voltage level and size, the effective
capacitance of COUT at target output level should meet the value in above table to make converter operated in
stable and normal.
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Typical Operating Characteristics
L : WE-74404054022 (DCR = 19m) for VOUT = 1V and 1.8V.
L : WE-74404054047 (DCR = 30m) for VOUT = 3.3V and 5V.
Efficiency vs. Output Current
Efficiency vs. Output Current
100
100
90
90
80
70
VOUT = 3.3V
60
VOUT = 1.8V
50
VOUT = 1V
Efficiency (%)
Efficiency (%)
80
40
30
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.8V
40
VOUT = 1V
30
20
20
10
0
0.001
70
10
RT6252AHGJ6F, VIN = 5V
0.01
0.1
1
0
0.001
10
RT6252AHGJ6F, VIN = 12V
0.01
90
90
80
80
VOUT = 3.3V
VOUT = 1.8V
50
VOUT = 1V
40
30
70
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.8V
40
VOUT = 1V
30
20
20
10
0
0.001
10
RT6252BHGJ6F, VIN = 5V
0.01
0.1
1
0
0.001
10
RT6252BHGJ6F, VIN = 12V
0.01
Efficiency vs. Output Current
100
90
90
10
80
70
VOUT = 3.3V
60
VOUT = 1.8V
Efficiency (%)
80
Efficiency (%)
1
Efficiency vs. Output Current
100
VOUT = 1V
50
40
30
70
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.8V
40
VOUT = 1V
30
20
20
10
10
RT6252AHGH6F, VIN = 5V
0.01
0.1
1
Output Current (A)
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DS6252A/B-02
0.1
Output Current (A)
Output Current (A)
0
0.001
10
Efficiency vs. Output Current
100
Efficiency (%)
Efficiency (%)
Efficiency vs. Output Current
100
60
1
Output Current (A)
Output Current (A)
70
0.1
August 2021
10
0
0.001
RT6252AHGH6F, VIN = 12V
0.01
0.1
1
10
Output Current (A)
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13
RT6252A/B
Efficiency vs. Output Current
100
90
90
80
80
70
Efficiency (%)
Efficiency (%)
Efficiency vs. Output Current
100
VOUT = 3.3V
60
VOUT = 1.8V
50
VOUT = 1V
40
30
70
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.8V
40
VOUT = 1V
30
20
20
10
10
RT6252BHGH6F, VIN = 5V
0
0.001
0.01
0.1
1
RT6252BHGH6F, VIN = 12V
0
0.001
10
0.01
0.1
1
10
Output Current (A)
Output Current (A)
Output Voltage vs. Output Current
Output Voltage vs. Output Current
1.20
5.50
1.10
Output Voltage (V)
Output Voltage (V)
1.15
1.05
1.00
RT6252AHGJ6F, VIN = 5V
0.95
RT6252BHGJ6F, VIN = 5V
0.90
RT6252AHGJ6F, VIN = 12V
0.85
RT6252BHGJ6F, VIN = 12V
5.25
5.00
RT6252AHGJ6F, VIN = 9V
RT6252BHGJ6F, VIN = 9V
4.75
RT6252AHGJ6F, VIN = 12V
RT6252BHGJ6F, VIN = 12V
VOUT = 5V
VOUT = 1V
4.50
0.80
0
0.5
1
1.5
0
2
0.5
1
1.5
2
Output Current (A)
Output Current (A)
Output Voltage vs. Output Current
Output Voltage vs. Output Current
1.20
5.50
1.10
Output Voltage (V)
Output Voltage (V)
1.15
1.05
1.00
RT6252AHGH6F, VIN = 5V
0.95
RT6252BHGH6F, VIN = 5V
0.90
RT6252AHGH6F, VIN = 12V
0.85
RT6252BHGH6F, VIN = 12V
5.25
5.00
RT6252AHGH6F, VIN = 9V
RT6252BHGH6F, VIN = 9V
4.75
RT6252AHGH6F, VIN = 12V
RT6252BHGH6F, VIN = 12V
VOUT = 5V
VOUT = 1V
4.50
0.80
0
0.5
1
1.5
Output Current (A)
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14
2
0
0.5
1
1.5
2
Output Current (A)
is a registered trademark of Richtek Technology Corporation.
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August 2021
RT6252A/B
Output Voltage vs. Input Voltage
Output Voltage vs. Input Voltage
1.10
5.5
5.3
1.05
1.00
Output Voltage (V)
Output Voltage (V)
5.4
RT6252AHGJ6F
RT6252BHGJ6F
0.95
5.2
5.1
RT6252AHGJ6F
RT6252BHGJ6F
5.0
4.9
4.8
4.7
4.6
VOUT = 1V
VOUT = 5V
4.5
0.90
5
6
7
8
9
9
10 11 12 13 14 15 16 17
10
11
12
13
14
15
16
17
Input Voltage (V)
Input Voltage (V)
Output Voltage vs. Input Voltage
Output Voltage vs. Input Voltage
1.10
5.5
5.3
1.05
1.00
Output Voltage (V)
Output Voltage (V)
5.4
RT6252AHGH6F
RT6252BHGH6F
0.95
5.2
5.1
5.0
RT6252AHGH6F
RT625BAHGH6F
4.9
4.8
4.7
4.6
VOUT = 1V
VOUT = 5V
4.5
0.90
5
6
7
8
9
9
10 11 12 13 14 15 16 17
10
11
12
Quiescent Current vs. Temperature
14
15
16
17
Shutdown Current vs. Temperature
340
1.0
RT6252AHGJ6F
RT6252AHGJ6F
Shutdown Current (μA)1
320
Quiescent Current (μA)
13
Input Voltage (V)
Input Voltage (V)
300
280
VIN = 17V
260
VIN = 12V
VIN = 9V
240
VIN = 5V
220
0.5
0.0
VIN = 17V
VIN = 12V
-0.5
VIN = 9V
VIN = 5V
-1.0
200
-50
-25
0
25
50
75
100
Temperature (°C)
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DS6252A/B-02
August 2021
125
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT6252A/B
Quiescent Current vs. Temperature
Shutdown Current vs. Temperature
340
1.0
RT6252AHGH6F
RT6252AHGH6F
Shutdown Current (μA)1
Quiescent Current (μA)
320
300
280
VIN = 17V
260
VIN = 12V
240
VIN = 9V
VIN = 5V
220
0.5
0.0
VIN = 17V
VIN = 12V
-0.5
VIN = 9V
VIN = 5V
-1.0
200
-50
-25
0
25
50
75
100
-50
125
-25
0
50
75
100
125
Reference Voltage vs. Temperature
0.85
0.8
0.84
0.79
0.83
Reference Voltage (V)
Reference Voltage (V)
Reference Voltage vs. Temperature
0.81
0.78
0.77
0.76
0.75
0.74
0.73
0.72
25
Temperature (°C)
Temperature (°C)
0.82
0.81
0.80
0.79
0.78
0.77
0.76
RT6252AHGJ6F
RT6252AHGH6F
0.75
0.71
-50
-25
0
25
50
75
100
-50
125
-25
0
25
50
75
100
125
Temperature (°C)
Temperature (°C)
Frequency vs. Input Voltage
Frequency vs. Output Current
700
700
600
Frequency (kHz)1
Frequency (kHz)1
650
600
550
500
400
300
200
500
100
450
5
6
7
8
9
10 11 12 13 14 15 16 17
Input Voltage (V)
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16
0
0.001
0.01
0.1
1
10
Output Current (A)
is a registered trademark of Richtek Technology Corporation.
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August 2021
RT6252A/B
EN Threshold vs. Temperature
1.4
650
1.3
EN Threshold (V)
Frequency (kHz)1
Frequency vs. Temperature
700
600
550
EN VIH
1.2
1.1
EN VIL
500
1.0
450
0.9
RT6252AHGJ6F
-50
-25
0
25
50
75
100
-50
125
-25
0
EN Threshold vs. Temperature
75
100
125
4.1
UVLO_H
4.0
EN VIH
3.9
1.3
Input Voltage (V)
EN Threshold (V)
50
UVLO vs. Temperature
1.4
1.2
1.1
EN VIL
3.8
3.7
3.6
UVLO_L
3.5
3.4
3.3
RT6252HGH6F
3.2
RT6252AHGJ6F
3.1
1.0
-50
-25
0
25
50
75
100
-50
125
UVLO vs. Temperature
4.3
-25
0
25
50
75
100
125
Temperature (°C)
Temperature (°C)
Power On from EN
UVLO_H
4.2
VOUT
(5V/Div)
4.1
Input Voltage (V)
25
Temperature (°C)
Temperature (°C)
4.0
VEN
(2V/Div)
3.9
VSW
(10V/Div)
3.8
UVLO_L
3.7
3.6
3.5
IL
(1A/Div)
RT6252AHGH6F
3.4
-50
-25
0
25
50
75
100
Temperature (°C)
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DS6252A/B-02
August 2021
125
VIN = 12V, VOUT = 5V
IOUT = 2A, L = 4.7H
Time (1ms/Div)
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RT6252A/B
Power Off from EN
VOUT
(5V/Div)
VOUT
(5V/Div)
VEN
(2V/Div)
VIN
(10V/Div)
VSW
(10V/Div)
IL
(1A/Div)
Power On from VIN
VSW
(10V/Div)
VIN = 12V, VOUT = 5V
IOUT = 2A, L = 4.7H
IL
(1A/Div)
Time (200s/Div)
Time (5ms/Div)
Power Off from VIN
Output Ripple as IOUT = 10mA
VOUT
(5V/Div)
VOUT
(50mV/Div)
VIN
(10V/Div)
VSW
(10V/Div)
IL
(1A/Div)
VSW
(5V/Div)
VIN = 12V, VOUT = 5V
IOUT = 2A, L = 4.7H
IL
(1A/Div)
VIN = 12V, VOUT = 5V, IOUT = 10mA, L = 4.7H
Time (5ms/Div)
Time (10s/Div)
Output Ripple as IOUT = 2A
Load Transient (No Load to Full Load)
VOUT
(50mV/Div)
VOUT
(20mV/Div)
VIN = 12V, VOUT = 5V
IOUT = 0A to 2A, L = 4.7H
VSW
(5V/Div)
IL
(1A/Div)
VIN = 12V, VOUT = 5V, IOUT = 2A, L = 4.7H
Time (1s/Div)
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18
VIN = 12V, VOUT = 5V
IOUT = 2A, L = 4.7H
IOUT
(1A/Div)
Time (400s/Div)
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RT6252A/B
Over-Current Protection and UVP
Load Transient (Half Load to Full Load)
VOUT
(2V/Div)
VOUT
(50mV/Div)
VIN = 12V, VOUT = 5V
IOUT = 1A to 2A, L = 4.7H
VIN = 12V, VOUT = 5V
L = 4.7H
VSW
(10V/Div)
IOUT
(500mA/Div)
IL
(2A/Div)
VOUT
(2V/Div)
Time (400s/Div)
Time (20s/Div)
Short Circuit Protection
Short Circuit before Power On
VIN = 12V, VOUT = 5V, L = 4.7H
VOUT
(5V/Div)
VIN
(10V/Div)
VSW
(10V/Div)
VSW
(10V/Div)
IL
(2A/Div)
IL
(2A/Div)
Time (10ms/Div)
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DS6252A/B-02
VIN = 12V, VOUT = 5V, L = 4.7H
August 2021
Time (10ms/Div)
is a registered trademark of Richtek Technology Corporation.
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19
RT6252A/B
Application Information
The output stage of a synchronous buck converter is
composed of an inductor and capacitor, which stores
and delivers energy to the load, and forms a
second-order low-pass filter to smooth out the switch
node voltage to maintain a regulated output voltage.
Inductor Selection
The inductor selection trade-offs among size, cost,
efficiency, and transient response requirements.
Generally, three key inductor parameters are specified
for operation with the device: inductance value (L),
inductor saturation current (ISAT), and DC resistance
(DCR).
A good compromise between size and loss is to choose
the peak-to-peak ripple current equals to 20% to 50%
of the IC rated current. The switching frequency, input
voltage, output voltage, and selected inductor ripple
current determines the inductor value as follows :
L=
VOUT VIN VOUT
VIN fSW IL
Once an inductor value is chosen, the ripple current
(IL) is calculated to determine the required peak
inductor current.
IL =
VOUT VIN VOUT
I
and IL(PEAK) = IOUT(MAX) L
VIN fSW L
2
IL(PEAK) should not exceed the minimum value of IC's
calculated inductance value is :
L
1.2 12 1.2
2.33μH
12 580kHz 0.8A
For the typical application, a standard inductance value
of 2.2H can be selected.
IL =
1.2 12 1.2
= 0.85A (42.5% of the IC rated current)
12 580kHz 2.2μH
and IL(PEAK) = 2A + 0.85A = 2.425A
2
For the 2.2H value, the inductor's saturation and
thermal rating should exceed at least 2.425A. For more
conservative, the rating for inductor saturation current
must be equal to or greater than switch current limit of
the device rather than the inductor peak current.
For EMI sensitive application, choosing shielding type
inductor is preferred.
Input Capacitor Selection
Input capacitance, CIN, is needed to filter the pulsating
current at the drain of the high-side power MOSFET.
CIN should be sized to do this without causing a large
variation in input voltage. The waveform of CIN ripple
voltage and ripple current are shown in Figure 1. The
peak-to-peak voltage ripple on input capacitor can be
estimated as the equation below :
through the inductor is the inductor ripple current plus
VCIN = D IOUT 1 D + IOUT ESR
CIN fSW
the output current. During power up, faults, or transient
where
upper current limit level. Besides, the current flowing
load conditions, the inductor current can increase above
the calculated peak inductor current level calculated
above. In transient conditions, the inductor current can
increase up to the switch current limit of the device. For
this reason, the most conservative approach is to
specify an inductor with a saturation current rating which
is equal to or greater than the switch current limit rather
than the peak inductor current.
Considering the Typical Application Circuit for 1.2V
output at 2A and an input voltage of 12V, using an
inductor ripple of 0.8A (40% of the IC rated current), the
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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20
D=
VOUT
VIN
For ceramic capacitors, the equivalent series
resistance (ESR) is very low, the ripple which is caused
by ESR can be ignored, and the minimum input
capacitance can be estimated as the equation below :
CIN_MIN = IOUT_MAX
D 1 D
VCIN_MAX fSW
where VCIN_MAX 200mV
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DS6252A/B-02
August 2021
RT6252A/B
capacitor should be 0402 or 0603 in size.
VCIN
CIN Ripple Voltage
VESR = IOUT x ESR
(1-D) x IOUT
CIN Ripple Current
D x IOUT
D x tSW (1-D) x tSW
Figure 1. CIN Ripple Voltage and Ripple Current
In addition, the input capacitor needs to have a very
low ESR and must be rated to handle the worst-case
RMS input current of :
IRMS IOUT_MAX
VOUT
VIN
1
VIN
VOUT
It is common to use the worse IRMS IOUT/2 at VIN =
2VOUT for design. Note that ripple current ratings from
capacitor manufacturers are often based on only 2000
hours of life which makes it advisable to further de-rate
the capacitor, or choose a capacitor rated at a higher
temperature than required.
Several capacitors may also be paralleled to meet size,
height and thermal requirements in the design. For low
input voltage applications, sufficient bulk input
capacitance is needed to minimize transient effects
during output load changes.
Ceramic capacitors are ideal for switching regulator
applications because of its small size, robustness, and
very low ESR. However, care must be taken when
these capacitors are used at the input. A ceramic input
capacitor combined with trace or cable inductance
forms a high quality (under damped) tank circuit. If the
RT6252A/B circuit is plugged into a live supply, the
Output Capacitor Selection
The RT6252A/B are optimized for ceramic output
capacitors and best performance will be obtained using
them. The total output capacitance value is usually
determined by the desired output voltage ripple level and
transient response requirements for sag (undershoot on
load apply) and soar (overshoot on load release).
Output Ripple
The output voltage ripple at the switching frequency is
a function of the inductor current ripple going through
the output capacitor’s impedance. To derive the output
voltage ripple, the output capacitor with capacitance
(COUT) and its equivalent series resistance (RESR)
must be taken into consideration. The output
peak-to-peak ripple voltage (VRIPPLE) caused by the
inductor current ripple (IL) is characterized by two
components, which are ESR ripple (VRIPPLE(ESR)) and
capacitive ripple (VRIPPLE(C)) and can be expressed as
below :
VRIPPLE = VRIPPLE(ESR) VRIPPLE(C)
VRIPPLE(ESR) = IL RESR
VRIPPLE(C) =
IL
8 COUT fSW
As ceramic capacitors are used, both parameters
should be estimated due to the extremely low ESR and
relatively small capacitance. Refer to the RT6252A/B's
typical application circuit of 1.2V application, the actual
inductor current ripple (IL) is 0.85A, and the output
capacitors are 2 x 22F (Murata ceramic capacitor:
GRM219R60J226ME47), VRIPPLE can be obtained as
below.
input voltage can ring to twice its nominal value,
possibly exceeding the device’s rating. This situation is
The ripple caused by ESR (2m) can be calculated
easily avoided by placing the low ESR ceramic input
capacitor in parallel with a bulk capacitor with higher
VRIPPLEESR = 0.85A 2m = 1.7mV
ESR to damp the voltage ringing.
Considering the capacitance derating, the effective
capacitance is approximately 18F as the output
The input capacitor should be placed as close as
possible to the VIN pins, with a low inductance
connection to the GND of the IC. In addition to a larger
bulk capacitor, a small ceramic capacitors of 0.1F
should be placed close to the VIN and GND pin. This
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6252A/B-02
August 2021
as :
voltage is 1.2V, and another parameter is :
0.85A
= 5.1mV
8 2 18μF 580kHz
= 1.7mV + 5.1mV = 6.8mV
VRIPPLE C =
VRIPPLE
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RT6252A/B
Output Transient Undershoot and Overshoot
In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load
steps up and down abruptly. The ACOT® transient
response is very quick and output transients are
usually small. The following section shows how to
calculate the worst-case voltage swings in response to
very fast load steps.
Both undershoot voltage and overshoot voltage consist
of two factors : the voltage steps caused by the output
capacitor's ESR, and the voltage sag and soar due to
the finite output capacitance and the inductor current
slew rate. Use the following formulas to check if the
ESR is low enough (typically not a problem with
ceramic capacitors) and the output capacitance is large
enough to prevent excessive sag and soar on very fast
load step edges, with the chosen inductor value.
load step, the output capacitor value, the inductor value
and the output voltage :
VSOAR =
Because some modern digital loads can exhibit nearly
instantaneous load changes, the amplitude of the ESR
should be taken into consideration while calculating the
VSAG & VSOAR.
Output Voltage Setting
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected
to FB, as shown in Figure 2. The output voltage is set
according to the following equation :
VOUT = 0.765V x (1 + RFB1 / RFB2)
VOUT
RFB1
The amplitude of the ESR step up or down is a function
of the load step and the ESR of the output capacitor :
FB
RT6252A/B
VESR _STEP = IOUT x RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,
the input-to-output voltage differential, and the
maximum duty cycle. The maximum duty cycle during a
fast transient is a function of the on-time and the
minimum off-time since the ACOT® control scheme will
ramp the current using on-times spaced apart with
L (IOUT )2
2 COUT VOUT
RFB2
GND
Figure 2. Output Voltage Setting
Place the FB resistors within 5mm of the FB pin.
Choose RFB2 between 10k and 100k to minimize
power consumption without excessive noise pick-up
and calculate RFB1 as follows :
RFB2 (VOUT VREF )
VREF
minimum off-times, which is as fast as allowed.
Calculate the approximate on-time (neglecting
RFB1
parasites) and maximum duty cycle for a given input
and output voltage as :
For output voltage accuracy, use divider resistors with
1% or better tolerance.
tON =
VOUT
tON
and DMAX =
VIN fSW
tON tOFF_MIN
The real on-time will slightly extend due to the voltage
drop which is related to output current; however, this
on-time compensation can be neglected. Besides, the
minimum on-time is 60ns, typ. If the calculated on-time
is smaller than minimum on-time, it and VOUT will both
be clamped. Calculate the output voltage sag as :
VSAG =
L (IOUT )2
2 COUT VIN(MIN) DMAX VOUT
The amplitude of the capacitive soar is a function of the
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22
Feed-Forward Capacitor Selection (CFF)
The RT6252A/B is optimized for low duty-cycle
applications and the control loop is stable with low ESR
ceramic output capacitors. In higher duty-cycle
applications (higher output voltages or lower input
voltages), the internal ripple signal will increase in
amplitude. Before the ACOT® control loop can react to
an output voltage fluctuation, the voltage change on the
feedback signal must exceed the internal ripple
amplitude. Because of the large internal ripple in this
condition, the response may become slower and
under-damped. This situation will result in ringing
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DS6252A/B-02
August 2021
RT6252A/B
waveform at output terminal. In case of high output
voltage application, the phenomenon described above
is more visible because of large attenuation in
inaccuracy. If the output voltage is over spec caused by
calculated CFF, please decrease the value of
feedforward capacitor CFF.
feedback network. As shown in Figure 3, adding a
feedforward capacitor (CFF) across the upper feedback
resistor is recommended. This increases the damping
of the control system.
L
VOUT
SW
RT6252A/B
RFB1
CFF
COUT
FB
GND
RFB2
Figure 3. Feedback Loop with Feedforward Capacitor
Loop stability can be checked by viewing the load
transient response. A load step with a speed that
exceeds the converter bandwidth must be applied. For
ACOT® , loop bandwidth can be in the order of 100 to
Figure 5. Load Transient Response With and Without
Feedforward Capacitor
200kHz, so a load step with 500ns maximum rising
time (dI/dt ≈ 2A/s) ensures the excitation frequency is
sufficient. It is important that the converter operates in
PWM mode, outside the light load efficiency range, and
below any current limit threshold. A load transient from
30% to 60% of maximum load is reasonable which is
Enable Operation
shown in Figure 4.
REN_DN, which is connected form EN to GND, ensures
that the chip still stays in shutdown even if EN pin is
floated.
60% Load
30% Load
Figure 4. Example of Measuring the Converter BW by
Fast Load Transient
CFF can be calculated basing on below equation :
1
2 BW
1 1 + 1
RFB1 RFB1 RFB2
Figure 5. shows the transient performance with and
without feedfoward capacitor.
Note that, after defining the CFF please also check the
load regulation, because feedforward capacitor might
inject an offset voltage into VOUT to cause VOUT
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6252A/B-02
rises above VUVLO while the EN pin voltage exceeds
VEN_H. The RT6252A/B is disabled when the VIN pin
voltage falls below VUVLO VUVLO or when the EN pin
voltage is below VEN_L. An internal pull-down resistor
For automatic start-up, the EN pin, with high-voltage
rating, can be connected to the input supply VIN directly
as shown in Figure 6.
fCO
CFF =
The RT6252A/B is enabled when the VIN pin voltage
August 2021
The built-in hysteresis band makes the EN pin useful
for simple delay and timing circuits. The EN pin can be
externally connected to VIN by adding a resistor REN
and a capacitor CEN, as shown in Figure 7, to have an
additional delay. The time delay can be calculated by
the equation below with the EN's internal threshold, at
which switching operation begins.
CEN =
t
Rth ln
Vth
Vth VEN_H
, where
Rth = REN // REN_DN
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23
RT6252A/B
Vth = VIN
REN_DN
REN_DN + REN
VIN
RT6252A/B
REN
An external MOSFET can be used for logic control
which is shown in Figure 8. In this case, REN is
connected between VIN and the EN pin. The MOSFET
Q1 will be under logic control to pull down the EN pin.
If the device is desired to be shut down by EN pin
before VIN falls below the UVLO threshold, a resistive
divider (REN1 and REN2) can be used to externally set
the input under-voltage lockout threshold as shown in
Figure 9. For a given REN1, REN2 can be found by the
equation below for the desired VIN stop voltage.
VIN_STOP
REN2 //REN_DN
< VEN_L
REN1 + REN2 //REN_DN
After REN1 and REN2 are defined, the input voltage
VIN_START is obtained from
VEN_H
REN1 + REN2 //REN_DN
= VIN_START
REN2 //REN_DN
VIN
RT6252A/B
REN_DN
Figure 6. Automatic Start-Up Setting
RT6252A/B
REN
EN
CEN
Enable
REN_DN
Q1
Figure 8. Digital Enable Control Circuit
RT6252A/B
VIN
REN1
EN
REN2
RENL_DN
Figure 9. Resistor Divider for Lockout Threshold
Setting
If VIN shuts down faster than VOUT and VOUT is larger
than 3.7V, buck converter becomes boost converter
and generates negative current. To prevent these
condition, EN should be shut down before VIN falls
below VOUT. Therefore, the resistor divider for lockout
threshold is recommended.
EN
VIN
EN
REN_DN
Figure 7. External Timing Control
Bootstrap Driver Supply
The bootstrap capacitor (CBOOT) between the BOOT
pin and the SW pin is used to create a voltage rail
above the applied input voltage, VIN. Specifically, the
bootstrap capacitor is charged through an internal
diode to a voltage equal to approximately PVCC each
time the low-side switch is turned on. The charge on
this capacitor is then used to supply the required
current during the remainder of the switching cycle. For
most applications, a 0.1F, 0603 ceramic capacitor
with X5R is recommended and the capacitor should
have a 6.3 V or higher voltage rating.
External Bootstrap Diode (Optional)
A bootstrap capacitor of 0.1F low-ESR ceramic
capacitor is connected between the BOOT and SW pins
to supply the high-side gate driver. It is recommended to
add an external bootstrap diode between an external 5V
voltage supply and the BOOT pin as shown in Figure 10
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is a registered trademark of Richtek Technology Corporation.
DS6252A/B-02
August 2021
RT6252A/B
to improve efficiency when the input voltage is below 5.5V.
The bootstrap diode can be a low-cost one, such as
1N4148 or BAT54. The external 5V can be a fixed 5V
voltage supply from the system, or a 5V output voltage
generated by the RT6252A/B. Note that the BOOT
voltage VBOOT must be lower than 5.5V.
RBOOT
BOOT
CBOOT
0.1μF
RT6252A/B
SW
Figure 11. External Bootstrap Resistor at the BOOT
Pin
5V
5V
DBOOT
DBOOT
BOOT
RT6252A/B
RBOOT
CBOOT
0.1μF
BOOT
SW
CBOOT
0.1μF
RT6252A/B
SW
Figure 10. External Bootstrap Diode
External Bootstrap Resistor (Optional)
Figure12. External Bootstrap Diode and Resistor at
the BOOT Pin
The gate driver of an internal power MOSFET, utilized
as a high-side switch, is optimized for turning on the
switch. The gate driver is not only fast enough for
reducing switching power loss, but also slow enough
for minimizing EMI. The EMI issue is worse when the
switch is turned on rapidly due to the induced high di/dt
noises. When the high-side switch is turned off, the
discharging time on SW node is relatively slow
because there’s the presence of dead time, both
high-side and low-side MOSFETs are turned off in this
interval. In some cases, it is desirable to reduce EMI
further, even at the expense of some additional power
dissipation. The turn-on rate of the high-side switch can
be slowed by placing a small bootstrap resistor RBOOT
between the BOOT pin and the external bootstrap
capacitor as shown in Figure 11. The recommended
range for the RBOOT is several ohms to 47 ohms, and it
could be 0402 or 0603 in size.
This will slow down the rates of the high-side switch
turn on and the rise of VSW . In order to improve EMI
performance and enhancement of the internal
MOSFET switch, the recommended application circuit
is shown in Figure 12, which includes an external
bootstrap diode for charging the bootstrap capacitor
and a bootstrap resistor RBOOT placed between the
BOOT pin and the capacitor/diode connection.
Thermal Considerations
In many applications, the RT6252A/B does not
generate much heat due to its high efficiency and low
thermal resistance of its TSOT-23-6 (FC) package.
However, in applications which the RT6252A/B runs at
a high ambient temperature and high input voltage, the
generated heat may exceed the maximum junction
temperature of the part.
The RT6252A/B includes an over-temperature protection
(OTP) circuitry to prevent overheating due to excessive
power dissipation. If the junction temperature reaches
approximately 155°C, the RT6252A/B stops switching
the power MOSFETs until the temperature is cooled
down by 35°C.
Note that the over-temperature protection is intended to
protect the device during momentary overload
conditions. The protection is activated outside of the
absolute maximum range of operation as a secondary
fail-safe and therefore should not be relied upon
operationally. Continuous operation above the
specified absolute maximum operating junction
temperature may impair device
permanently damage the device.
DS6252A/B-02
August 2021
or
The maximum power dissipation can be calculated by
the following formula :
PDMAX = TJMAX TA
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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/ θJAEFFECTIVE
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RT6252A/B
temperature is 125°C. TA is the ambient operating
temperature, and θJA(EFFECTIVE) is the system-level
junction to ambient thermal resistance. It can be
estimated from thermal modeling or measurements in
the system.
The thermal resistance of the device strongly depends
on the surrounding PCB layout and can be improved by
providing a heat sink of surrounding copper ground.
The addition of backside copper with thermal vias,
stiffeners, and other enhancements can also help
reduce thermal resistance.
As an example, considering the RT6252A is used in
application where VIN = 12V, IOUT = 2A, f SW = 580kHz,
VOUT = 5V. The efficiency at 5V, 2A is 91.3% by using
WE-74404054047 (4.7H, 30m DCR) as the inductor
and measured at room temperature. The core loss can
be obtained from its website, and it's 131mW. In this
case, the power dissipation of the RT6252A is
PD, RT =
1 η
2
POUT IO
DCR + PCORE
η
= 0.702W
Considering the system-level θJA(EFFECTIVE) is 69.6°C /W
(other heat sources are also considered), the junction
temperature of the regulator operating in a 25°C ambient
temperature is approximately :
The external
power dissipation caused by the
increasing RDS(ON) at higher temperature can be
calculated as
PD,RDS ON = 2A
2
5
5
2
20m + 2A 1
9m = 0.054W
12
12
As a result, the new power dissipation is 0.756W due to
the variation of RDS(ON). Therefore, the estimated new
junction temperature is
TJ' = 0.756W 69.6C/W + 60C = 112.6C
If the application requires a higher ambient temperature
and may exceed the recommended maximum junction
temperature of 125°C, care should be taken to reduce
the temperature rise of the part by using a heat sink or
air flow.
Resistance vs. Temperature
250
200
Resistance (mΩ)
where TJ(MAX) is the maximum allowed junction
temperature of the die. For recommended operating
condition specifications, the maximum junction
RDS(ON)_H
150
RDS(ON)_L
100
50
TJ = 0.63W 76C/W + 25C = 73.9C
0
Figure 13 shows the RT6252A/B RDS(ON) versus
different junction temperatures. If the application
requires a higher ambient temperature, the device
power dissipation and the junction temperature of the
device need to be recalculated based on a higher
RDS(ON) since it increases with temperature.
-50
-25
0
25
50
75
100
125
Temperature (°C)
Figure 13. RT6252A/B RDS(ON) vs. Temperature
Using 60°C ambient temperature as an example. Due
to the variation of junction temperature is dominated by
the ambient temperature, the T'J at 60°C ambient
temperature can be pre-estimated as
TJ' = 73.9C + 60C 25C = 108.9C
According to Figure 13, the increasing RDS(ON) can be
found as
RDS ON _H = 190m (at 108.9C) 170m 73.9C = 20m
RDS ON _L = 101m (at 108.9C) 92m 73.9C = 9m
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is a registered trademark of Richtek Technology Corporation.
DS6252A/B-02
August 2021
RT6252A/B
Layout Considerations
Follow the PCB layout guidelines below for optimal
performance of the device.
Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable
and jitter-free operation. The high current path
comprising of input capacitor, high-side FET,
inductor, and the output capacitor should be as short
as possible. This practice is essential for high
efficiency.
Place the input MLCC capacitors as close to the VIN
and GND pins as possible. The major MLCC
capacitors should be placed on the same layer as
the RT6252A/B.
SW node is with high frequency voltage swing and
should be kept at small area. Keep analog
components away from the SW node to prevent
stray capacitive noise pickup.
Connect feedback network behind the output
capacitors. Place the feedback components next to
the FB pin.
For better thermal performance, design a wide and
thick plane for GND pin or add a lot of vias to GND
plane.
An example of PCB layout guide is shown in Figure 14.
Add extra vias for thermal dissipation
Keep the SW node at small
area and keep analog
components away from the
SW node to prevent stray
capacitive noise pickup.
GND
GND
CBOOT RBOOT
SW
SW
1
REN
VIN
The VIN trace should
have enough width,
and use several vias to
shunt the high input
current.
L
CIN3
CIN2
Place the input MLCC capacitors as
close to the VIN and GND pins as
possible.
COUT1
Connect feedback network
behind the output
Place the feedback
components next to the FB pin.
RFB1
RFB2
CIN1
VOUT
COUT2
CFF
VOUT
GND
GND
Figure 14. Layout Guide
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August 2021
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27
RT6252A/B
Outline Dimension
Dimensions In Millimeters
Symbol
Dimensions In Inches
Min.
Max.
Min.
Max.
A
0.700
1.000
0.028
0.039
A1
0.000
0.100
0.000
0.004
B
1.397
1.803
0.055
0.071
b
0.300
0.559
0.012
0.022
C
2.591
3.000
0.102
0.118
D
2.692
3.099
0.106
0.122
e
0.950
0.037
H
0.080
0.254
0.003
0.010
L
0.300
0.610
0.012
0.024
TSOT-23-6 (FC) Surface Mount Package
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is a registered trademark of Richtek Technology Corporation.
DS6252A/B-02
August 2021
RT6252A/B
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min
Max
Min
Max
A
0.500
0.600
0.020
0.024
A1
0.000
0.050
0.000
0.002
A3
0.080
0.180
0.003
0.007
b
0.150
0.300
0.006
0.012
D
1.500
1.700
0.059
0.067
E
1.500
1.700
0.059
0.067
E1
1.100
1.300
0.043
0.051
e
0.500
0.020
L
0.100
0.300
0.004
0.012
L1
0.200
0.400
0.008
0.016
SOT-563 (FC) Surface Mount Package
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DS6252A/B-02
August 2021
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29
RT6252A/B
Footprint Information
Package
TSOT-26/TSOT-26(FC)/SOT-26/SOT-26(COL)
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30
Footprint Dimension (mm)
Number of
Pin
P1
A
B
C
D
M
6
0.95
3.60
1.60
1.00
0.70
2.60
Tolerance
±0.10
is a registered trademark of Richtek Technology Corporation.
DS6252A/B-02
August 2021
RT6252A/B
Package
SOT-563(FC)
Footprint Dimension (mm)
Number of
Pin
P1
A
B
C
D
M
6
0.50
2.42
1.02
0.70
0.30
1.30
Tolerance
±0.10
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume
responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and
reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may
result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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August 2021
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