AN3286
Application note
EVLVIPER25L-10WSB: 5 V/10 W, quasi-resonant isolated flyback
Introduction
This document describes a 5 V - 2 A SMPS using VIPer25, a current-mode offline converter,
specifically designed to build quasi-resonant flyback converters. Quasi-resonant operation,
in conjunction with other features of the device (very low quiescent currents; valley skipping
operation at light load; burst mode operation at very low or disconnected load) helps to
reach very good requirements both in terms of efficiency and in terms of consumption from
the mains at light load, allowing to meet the most modern SMPS standards.
The other features of the device are: 800 V avalanche rugged power section, limiting current
with adjustable set point, onboard soft-start, safe auto-restart after a fault condition, and
feed-forward correction.
The available protections are: adjustable and accurate overvoltage protection, thermal
shutdown with hysteresis, and delayed overload protection.
Figure 1.
August 2011
Demonstration board image
Doc ID 17998 Rev 1
1/42
www.st.com
Contents
AN3286
Contents
1
2
3
Adapter features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.1
Circuit description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.2
Transformer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Testing the board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
2.1
Typical board waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
2.2
Regulation precision and output voltage ripple . . . . . . . . . . . . . . . . . . . . 14
2.3
Burst mode and output voltage ripple . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
2.4
Efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
2.5
Light load performances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.6
Overload protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
2.7
2nd OCP protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
2.8
Voltage feed-forward function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
2.9
Output overvoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
2.10
Brownout protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
2.11
EMI measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
2.12
Board layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Appendix A Test equipment and measurement of efficiency and light load
performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
2/42
Doc ID 17998 Rev 1
AN3286
List of tables
List of tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
Table 8.
Table 9.
Table 10.
Table 11.
Table 12.
Table 13.
Table 14.
Table 15.
Table 16.
Table 17.
Table 18.
Table 19.
Electrical specifications. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Bill of material . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1338.0019 transformer characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Transformer pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output voltage and VDD line-load regulation. . . . . . . . . . . . . . . . . . . . . . . . . .
Output voltage ripple. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Burst mode related output voltage ripple . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Active mode efficiencies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Line voltage averaged efficiency vs. load . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Energy efficiency criteria for standard models . . . . . . . . . . . . . . . . . . . . . . . .
Energy efficiency criteria for low voltage models . . . . . . . . . . . . . . . . . . . . . .
No load input power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Energy consumption criteria for no load . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Light load performance POUT = 25 mW . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Light load performance. POUT = 50 mW . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Converter efficiency when the input power is 1 W . . . . . . . . . . . . . . . . . . . . .
IOUT_OLP and IOUT_REST values . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Doc ID 17998 Rev 1
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List of figures
AN3286
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Figure 33.
Figure 34.
Figure 35.
Figure 36.
Figure 37.
Figure 38.
Figure 39.
Figure 40.
Figure 41.
Figure 42.
Figure 43.
Figure 44.
Figure 45.
Figure 46.
Figure 47.
Figure 48.
4/42
Demonstration board image . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Application schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Transformer size and pin diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Transformer size . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Drain current and voltage at max. load 115 VAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Drain current and voltage at max. load 230 VAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Drain current and voltage at max. load 90 VAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Drain current and voltage at max. load 265 VAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Drain current and voltage at 230 VAC 0.7 A (valley skip.). . . . . . . . . . . . . . . . . . . . . . . . . . 13
Drain current and voltage at 115 VAC 0.6 A (uneven cycles) . . . . . . . . . . . . . . . . . . . . . . . 13
Switching frequency vs. transformer input power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Output voltage ripple at 115 VAC - full load. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Output voltage ripple at 230 VAC - full load. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Output voltage ripple at 115 VAC - no load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
VFBbm, VFBbmhys at 115 VAC - no load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Output voltage ripple at 115 VAC - 50 mA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Output voltage ripple at 230 VAC - 50 mA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Efficiency vs. VIN_AC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Efficiency vs. load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Active mode efficiency vs. VIN_AC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Input voltage averaged efficiency vs. load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
ENERGY STAR efficiency criteria . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Input power vs. input voltage for no load and light load conditions . . . . . . . . . . . . . . . . . . 23
Converter efficiency when the input power is 1 W . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
OLP: output short and protection tripping . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
OLP: output short and protection tripping - zoom . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
OLP: steady-state (auto-restart mode) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
OLP: steady-state (auto-restart mode) - zoom . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
OLP: short removal and restart. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
OLP: short removal and restart - zoom. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
2nd OCP protection tripping . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Operating with secondary winding shorted - restart mode . . . . . . . . . . . . . . . . . . . . . . . . . 27
Operating with secondary winding shorted - restart mode – zoom1. . . . . . . . . . . . . . . . . . 28
Operating with secondary winding shorted - restart mode – zoom2. . . . . . . . . . . . . . . . . . 28
Implementing input voltage feed-forward function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
IOUT_OLP and IOUT_REST vs. VIN_AC diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Output overvoltage protection at 115 VAC 0.2 A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Output overvoltage protection at 115 VAC 0.2 A - zoom . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Auto-restart mode of the overvoltage protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Auto-restart mode of the overvoltage protection - zoom. . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Brownout function block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Input voltage steps from 60 VDC to 100 VDC - no turn-on. . . . . . . . . . . . . . . . . . . . . . . . . . 33
Input voltage steps from 60 VDC to 115 VDC - turn-on . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Input voltage steps from 115 VDC to 88 VDC - no turn-off. . . . . . . . . . . . . . . . . . . . . . . . . . 33
Input voltage steps from 115 VDC to 80 VDC - turn-off . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Quasi-peak measurement at 230 VAC - full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Average measurement at 230 VAC - full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Complete layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Doc ID 17998 Rev 1
AN3286
Figure 49.
Figure 50.
Figure 51.
Figure 52.
Figure 53.
List of figures
Top overlay . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Bottom layer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Connection of the UUT to the wattmeter for power measurements . . . . . . . .
Suggested connection for low power measurements (switch in position 1) . .
Suggested connection for high power measurements (switch in position 2) .
Doc ID 17998 Rev 1
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5/42
Adapter features
1
AN3286
Adapter features
Table 1 lists the electrical specifications of the demonstration board.
Table 1.
Electrical specifications
Parameter
Symbol
Value
VIN_AC
[90 VAC - 265 VAC]
VOUT
5V
IOUTmax
2A
Precision of output regulation
ΔVOUT_LF
±5%
High frequency output voltage ripple
ΔVOUT_HF
50 mV
Max. ambient operating temperature
TA
60 °C
Input voltage range
Output voltage
Max. output current
1.1
Circuit description
The power supply is set in a flyback topology and its schematic is shown in Figure 2. The
input section includes protection elements (fuse and NTC for inrush current limiting), a filter
for EMC suppression (C1, T2, C13), a diode bridge (BR1), and an electrolytic bulk capacitor
(C3) as the front-end AC-DC converter. The transformer uses a standard E25 ferrite core.
A Transil™ clamp network is used for leakage inductance demagnetization.
At power-up the DRAIN pin supplies the internal HV startup current generator which
charges the C4 capacitor up to VDDon. At this point the power MOSFET starts switching, the
generator is turned off, and the IC is powered by the energy stored in C4, until the auxiliary
winding voltage becomes high enough to sustain the operation through D1 and R1. The
value of the resistor R3 between the ZCD and GND pins defines the VIPer25 current limit
and, in conjunction with D2 and R14, also realizes the overvoltage protection function.
The ZCD pin is also responsible for the quasi-resonant operation, being a transformer
demagnetization sensing input triggering the MOSFET turn-on.
Converter power capability variations with the mains voltage, inherent in quasi-resonant
operation, are compensated by the R3-R15 voltage divider, which implements the line
voltage feed-forward correction function. At very light or disconnected load the device enters
burst mode, allowing to keep low the consumption from the mains in such conditions
(< 50 mW at 265 VAC). The output rectifier D4 has been selected according to the calculated
maximum reverse voltage, forward voltage drop, and power dissipation, and is a power
Schottky type. The output voltage regulation is performed by secondary feedback with
a TS431 driving an optocoupler, in this case, a PC817, ensuring the required insulation
between primary and secondary. The optotransistor drives directly the FB pin of the
VIPer25, which is connected to the compensation network made up of C6, C7, and R12.
A small LC network has been added at the output in order to filter the high frequency ripple
without increasing the output capacitor size and a 100 nF capacitor has been placed very
close to the output connector solder points, to limit the spike amplitude. Through jumper J,
the BR pin can be either connected to a voltage divider sensing the mains rectified voltage
in order to realize a brownout protection, or to GND. In the latter case, the RH1-RH2-RL
branch can be disconnected from the mains in order to reach minimum standby
consumption.
6/42
Doc ID 17998 Rev 1
Doc ID 17998 Rev 1
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AN3286
Adapter features
Application schematic
7/42
Adapter features
Table 2.
8/42
AN3286
Bill of material
Reference
Part
Description
BR1
DF06
C3
22 µF, 450 V
Electrolytic capacitor
C4
22 µF, 35 V
Electrolytic capacitor
C1, C13
100 nF
C5
N.C.
C6
2.2 nF, 50 V
Ceramic capacitor
C7
22 nF, 50 V
Polypropylene capacitor
C8
2.2 nF
C9
N.C.
Electrolytic capacitor
C10
100 µF, 25 V
Electrolytic capacitor
C14, C15
680 µF, 16 V
Ultra-low ESR electrolytic capacitor, MCZ
series
C11
27 nF, 50 V
Ceramic capacitor
C16
100 nF, 50 V
Ceramic capacitor
CL
10 nF, 50 V
Ceramic capacitor
D1
BAT46
Small signal diode
D2
1N4148
Small signal diode
D3
STTH1L06
Clamp diode
STMicroelectronics
D4
STPS5L60
Power diode
STMicroelectronics
D5
1.5KE250
Transil
STMicroelectronics
Diode bridge
Manufacturer
VISHAY®
X2 capacitor
Polypropylene capacitor
Y1 capacitor
Dz
18 V Zener diode
F1
Fuse 1.6 A 250 V
NTC1
2.2 Ω
Thermistor
R1
4.7 Ω
1/4 W resistor
R3
22 kΩ
1/4 W resistor
R6
15 kΩ
1/4 W resistor
R8
82 kΩ
1/4 W resistor, 1%
R9
27 kΩ
1/4 W resistor, 1%
R10
120 kΩ
1/4 W resistor
R12
56 kΩ
1/4 W resistor
R13
3.3 kΩ
1/4 W resistor
R14
68 kΩ
1/4 W resistor
R15
560 kΩ
1/4 W resistor
Rz
560 Ω
1/4 W resistor
RH1
1.1 MΩ
1/4 W resistor
Doc ID 17998 Rev 1
Rubycon
STMicroelectronics™
EPCOS
AN3286
Adapter features
Table 2.
Bill of material (continued)
Reference
Part
RH2
1.1 MΩ
1/4 W resistor
RL
12 kΩ
1/4 W resistor
VR1
TS431
Voltage reference
OPTO1
PC817
Optocoupler
T1
1338.0019
Transformer
Magnetica®
T2
4530R4B
Common mode choke
TDK
L1
2.2 µH
RFB0807-2R2L
Coilcraft
J
IC
1.2
Description
Manufacturer
STMicroelectronics
3-pin jumper
VIPer25
PWM controller
STMicroelectronics
Transformer
The transformer’s electrical characteristics are listed in Table 3.
Table 3.
1338.0019 transformer characteristics
Properties
Value
Test condition
Primary inductance
1.5 mH ±15%
Measured at 1 kHz
Leakage inductance
0.8% nom.
Measured at 10 kHz
Primary to secondary turn ratio (4 - 5)/(6, 7 - 10, 9) 12.85 ±5%
Measured at 10 kHz
Primary to auxiliary turn ratio (4 - 5)/(1 - 2)
5.29 ±5%
Measured at 10 kHz
Nominal operating frequency
60 kHz
VIPer25 demonstration board
Nominal/peak power
10 W/15 W
Saturation current
1A
BSAT = 0.32T
Insulation
4 kV
Primary to secondary
Doc ID 17998 Rev 1
9/42
Adapter features
AN3286
The size, the pinout, and the mechanical characteristics are given in Figure 3 and 4:
Figure 3.
Transformer size and pin diagram
-)33).'
0).2%&%2%.#%
!30#"
!33%-",).'
A "OTTOMVIEWPINSIDE
B %LECTRICALDIAGRAM
!-
Figure 4.
Transformer size
MAX
MAX
,ABEL
MAX
A 3IDEVIEW
A 3IDEVIEW
!-
Table 4.
10/42
Transformer pin description
Pin
Description
Pin
Description
5
Primary, to the DC input voltage (400 V)
6
4
Primary, to the drain of the MOSFET
7
Secondary output
5 V 2 A (3 Apk)
3
Removed
8
N. C.
2
Auxiliary GND
9
1
Auxiliary output
10
Secondary
GND
Doc ID 17998 Rev 1
AN3286
Testing the board
2
Testing the board
2.1
Typical board waveforms
The VIPer25 is operated in quasi-resonant mode. The ZCD (zero current detect) pin is able
to sense the transformer demagnetization and switch on the MOSFET on the valley of the
drain voltage ringing that follows the transformer demagnetization.
The converter has been designed so that in full load condition the MOSFET is always
switched on, on the first valley of the transformer demagnetization, within the entire input
voltage range. This operation mode is referred to as “quasi-resonant”, or “boundary”
between discontinuous conduction mode (DCM) and continuous conduction mode (CCM).
Figure 5 and 6 show drain current and voltage waveforms at nominal input voltages
(115 VAC and 230 VAC) and maximum load (2 A). Figure 7 and 8 show the same waveforms
in the same load condition, but at minimum and maximum input voltage respectively.
Figure 5.
Drain current and voltage at max.
load 115 VAC
Figure 6.
Doc ID 17998 Rev 1
Drain current and voltage at max.
load 230 VAC
11/42
Testing the board
Figure 7.
AN3286
Drain current and voltage at max.
load 90 VAC
Figure 8.
Drain current and voltage at max.
load 265 VAC
In quasi-resonant operation the switching frequency depends on the input/output conditions,
increasing with increasing mains voltage and with decreasing output load. In fact, as the
input voltage increases, the ON time is reduced and so the switching frequency increases;
as the load decreases, the drain peak current (then also the needed ON time and the
demagnetization time) decreases, therefore, the switching frequency increases.
In order to avoid the frequency increasing too much at medium/light load (which would result
in an increase of all the frequency-related losses), the device is provided with a frequency
fold-back function with an internal limit of 136 kHz (typ.); as the converter switching
frequency approaches the limit, this function acts by inhibiting the MOSFET switch-on, on
the first valley, and allowing it on the second, or the third and so on, depending on the
working condition, as shown in Figure 9 and 10 below. This mode of operation is referred to
as “valley skipping mode”.
In Figure 9, one ringing cycle of the drain voltage is always skipped. In this figure, what
should be the switching frequency without the frequency fold-back function (fQR), the
MOSFET turn-on inhibit time (1/fFF), and the actual converter switching frequency
(fValley_Skipping), is highlighted.
12/42
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Figure 9.
Testing the board
Drain current and voltage at
230 VAC 0.7 A (valley skip.)
Figure 10. Drain current and voltage at
115 VAC 0.6 A (uneven cycles)
While the frequency fold-back is active, uneven switching cycles may be observed, due to
the fact that the OFF time of the MOSFET is allowed to change with discrete steps
(MOSFET is switched on always in the valley) while the OFF time needed for cycle-by-cycle
energy balance may fall in between. One or more longer switching cycles is then
compensated by one or more shorter ones, and vice versa.
This phenomenon (shown in Figure 10) is completely normal and there is no appreciable
effect on the performance of the converter and its output voltage.
Figure 11 shows the changes of the switching frequency vs. the transformer input power
(almost proportional to the output power) for two different input voltages.
Figure 11. Switching frequency vs. transformer input power
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2.2
AN3286
Regulation precision and output voltage ripple
The output voltage of the board and the VDD pin voltage (rectified auxiliary output) have
been measured in different line and load conditions. The results are given in Table 5. The
output voltage practically is not affected by the line condition.
The VDD voltage increases with the load on the regulated output; in order to avoid it
exceeding its operating range an external clamp has been used (Dz, Rz).
Table 5.
Output voltage and VDD line-load regulation
No load
Half load
Full load
VIN (VAC)
VOUT (V)
VDD (V)
VOUT (V)
VDD (V)
VOUT (V)
VDD (V)
90
5.02
8.4
4.98
18.5
4.98
20.2
115
5.02
8.3
4.97
18.5
4.97
20.2
230
5.02
8.2
4.97
18.2
4.97
20.5
265
5.03
8.1
4.96
18.1
4.96
20.5
The ripple at the switching frequency superimposed at the output voltage has also been
measured and the results are reported in Table 6.
Table 6.
Output voltage ripple
1/10 load
Half load
Full load
VOUT (mV)
VOUT (mV)
VOUT (mV)
90
45
19
50
115
46
16
38
230
48
22
28
265
48
25
27
VIN (VAC)
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Testing the board
Figure 12. Output voltage ripple at 115 VAC full load
2.3
Figure 13. Output voltage ripple at 230 VAC full load
Burst mode and output voltage ripple
When the load is so low that the voltage at the FB pin falls below the VFBbm internal
threshold (0.6 V typical), the VIPer25 is disabled. At this point, the feedback reaction to the
stop of the energy delivery makes the FB pin voltage increase again, and when it goes
100 mV above the VFBbm threshold, the device restarts switching.
This results in a controlled on/off operation which is referred to as “burst mode” and is
shown in the figures below, which report the output voltage ripple, the FB pin voltage, and
the drain peak current when the converter is not or only lightly loaded and supplied with
115 VAC and 230 VAC respectively. During burst mode, the peak drain current value is ID_BM,
whose typical value is 160 mA, as shown in Figure 15. This mode of operation keeps the
frequency-related losses low when the load is very light or disconnected and makes it easier
to comply with energy saving regulations.
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Figure 14. Output voltage ripple at 115 VAC no load
Figure 15. VFBbm, VFBbmhys at 115 VAC - no
load
Figure 16. Output voltage ripple at 115 VAC 50 mA
Figure 17. Output voltage ripple at 230 VAC 50 mA
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Table 7 shows the measured value of the burst mode frequency ripple measured in different
operating conditions. The ripple in burst mode operation is very low and always below
50 mV.
Table 7.
2.4
Burst mode related output voltage ripple
VIN (VAC)
No load (mV)
25 mA load (mV)
50 mA load (mV)
90
24
34
38
115
23
33
37
230
22
32
38
265
21
35
39
Efficiency
This section and the following section report the results of efficiency and light load
measurements.
Appendix A at the end of this document provides some details on the settings of the power
measurement equipment.
According to the ENERGY STAR ® average active mode efficiency testing method, the
efficiency measurements have been done at full load and at 75%, 50%, and 25% of full load
for different input voltages. The results are reported in Table 8.
Table 8.
Efficiency
Efficiency (%)
VIN (VAC)
Full load (2 A)
75% load (1.5 A)
50% load (1 A)
25% load (0.5 A)
90
78.00
80.60
81.33
82.83
115
80.42
82.40
82.37
82.83
150
81.95
83.16
82.46
82.34
180
82.49
83.33
82.15
81.62
230
82.63
83.00
81.17
80.33
265
82.25
82.29
80.00
79.28
For better visibility, the results have also been plotted in the diagrams below. Figure 18 plots
the efficiency vs. VIN for the four different load values. Figure 19 reports the efficiency as
a function of the load for different input voltage values.
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Figure 18. Efficiency vs. VIN_AC
Figure 19. Efficiency vs. load
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The active mode efficiency is defined as the average of the efficiencies measured at 25%,
50%, and 75% of maximum load and at maximum load itself. Table 9 reports the active
mode efficiency calculated from the values in Table 8. For better visibility, Table 9 values are
plotted in Figure 20.
Table 9.
Active mode efficiencies
Active mode efficiency
VIN (VAC)
Efficiency (%)
90
80.67
115
82.00
150
82.48
180
82.40
230
81.78
265
80.97
Figure 20. Active mode efficiency vs. VIN_AC
In Table 10 and Figure 21 the averaged value of the efficiency versus load are reported
(average has been done considering the efficiency at different values of the input voltage).
Table 10.
Line voltage averaged efficiency vs. load
Load (% of full load)
Efficiency (%)
100
80.93
75
81.28
50
81.02
25
82.06
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Figure 21. Input voltage averaged efficiency vs. load
In version 2.0 of the ENERGY STAR program requirements for single voltage external ACDC and AC-AC power supplies (1.), the power supplies are divided into two categories: low
voltage power supplies and standard power supplies, with respect to the nameplate output
voltage and current. To be considered a low-voltage power supply, an external power supply
must have a nameplate output voltage of less than 6 V and a nameplate output current of
more than or equal to 550 mA.
The following tables report the EPA energy efficiency criteria for AC-DC power supplies in
active mode for standard models and low voltage models respectively.
Table 11.
Energy efficiency criteria for standard models
Nameplate output power (Pno)
Table 12.
20/42
Minimum average efficiency in active mode
(expressed as a decimal)
0 to ≤ 1 watt
≥ 0.48*Pno+0.140
> 1 to ≤ 49 watts
≥ [0.0626 * In (Pno)] + 0.622
> 49 watts
≥ 0.870
Energy efficiency criteria for low voltage models
Nameplate output power (Pno)
Minimum average efficiency in active mode
(expressed as a decimal)
0 to ≤ 1 watt
≥ 0.497 *Pno+0.067
> 1 to ≤ 49 watts
≥ [0.075 * In (Pno)] + 0.561
> 49 watts
≥ 0.860
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Testing the board
The criteria are plotted in Figure 22, where the red line depicts the criteria for the standard
model and the blue line the criteria for the low voltage model. The PNO axis uses
a logarithmic scale.
Figure 22. ENERGY STAR efficiency criteria
The presented power supply belongs to the low-voltage power supply category and, in order
to be compliant with ENERGY STAR requirements, needs to have an efficiency higher than
73.37%. For all the considered input voltages, the efficiency results (see Table 8) are higher
than the recommended value.
2.5
Light load performances
The input power of the converter was measured in no load conditions for different input
voltages. The results are given in Table 13.
Table 13.
No load input power
VIN (VAC)
PIN (mW)
90
15
115
17
150
20
180
23
230
28
265
33
Version 2.0 of the ENERGY STAR program also takes into consideration the power
consumption of the power supply when it is not loaded. The criteria to be compliant with are
reported in Table 14.
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Table 14.
Energy consumption criteria for no load
Nameplate output power (Pno)
Maximum power in no load for AC-DC EPS
0 to ≤ 50 watts
< 0.3 watts
> 50 watts < 250 watts
< 0.5 watts
The performance of the presented board is much better than required; the power
consumption is about ten times lower than the ENERGY STAR limit. Even if the
performance seems to be unproportionally better than required it is worth noting that often
AC-DC adapter or battery charger manufacturers have very strict requirements about no
load consumption, and if the converter is used as an auxiliary power supply, the line filter is
often the big line filter of the entire power supply which significantly increases consumption
in standby mode.
Even if the ENERGY STAR program does not have other requirements regarding light load
performance, in order to provide complete information the input power and efficiency of the
demonstration board, also in two other light load cases, is reported. Table 15 and 16 show
the board performances when the output load is 25 mW and 50 mW respectively.
Table 15.
VIN (VAC)
POUT (mW)
PIN (mW)
Eff. (%)
PIN - POUT (mW)
90
25
46
54.35
21
115
25
48
52.08
22
150
25
51
49.01
26
180
25
53
47.17
28
230
25
60
41.67
35
265
25
66
37.88
38
Table 16.
22/42
Light load performance POUT = 25 mW
Light load performance. POUT = 50 mW
VIN (VAC)
POUT (mW)
PIN (mW)
Eff. (%)
PIN - POUT (mW)
90
50
76.5
65.40
27
115
50
78
64.10
28
150
50
81
61.73
31
180
50
85
58.82
35
230
50
92
54.34
42
265
50
98
51.02
48
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Testing the board
Figure 23 shows the input power vs. input voltage for no load and light load conditions.
Figure 23. Input power vs. input voltage for no load and light load conditions
Depending on the equipment supplied, there are several criteria to measure the standby or
light load performance of a converter. One of these is the measurement of the output power
when the input power is equal to one watt. Table 17 shows the output power needed to
obtain 1 W of input power in different line conditions. Figure 24 shows the results of this
measurement.
Table 17.
Converter efficiency when the input power is 1 W
VIN (VAC)
PIN (W)
Eff. (%)
90
1
82.21
115
1
81.54
150
1
80.63
180
1
79.66
230
1
78.10
265
1
76.82
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Figure 24. Converter efficiency when the input power is 1 W
With the demonstration board presented here it is possible to have or to not have brownout
protection. All the efficiency and light load performances have been measured with
brownout protection disabled (series RH1, RH2, and RL in Figure 2 disconnected). If
brownout protection is required, for the same load condition, the input power increases,
because the brownout network is connected in parallel with the bulk capacitor (C3 in the
schematic) and dissipates some power according to the following formula:
Equation 1
P BR
LOSS
V
2
BULK RMS
= ---------------------------------------------RH1 + RH2 + RL
In light load condition the voltage across the bulk capacitor can be considered constant and
equal to the peak of the AC input voltage. Considering the worst case of maximum input
voltage the dissipation across the brownout network is:
Equation 2
P BR
LOSS
V
2
2
ACMAX PK
265 ⋅ 2 ) - = 64mW
= ---------------------------------------------- = (------------------------------6
RH1 + RH2 + RL
2.268 ⋅ 10
which is negligible when measuring the efficiency in not too light load condition but becomes
quite important when measuring the standby or light load performance.
2.6
Overload protection
The VIPer25 is a current mode converter. This means that the regulation of the output
voltage is made by increasing or decreasing the primary peak current on a cycle-by-cycle
basis as a consequence of the increase or decrease of the output power demand.
The peak current is internally sensed and converted into a voltage which is compared with
the FB pin voltage. The device is shut down as soon as the two voltages are equal.
When the FB pin voltage reaches VFBlin (3.3 V typical), the drain peak current reaches its
maximum value, IDLIM (which is 0.7 A typical or a lower value according to the value of the
resistor if connected between the ZCD and GND pins).
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If the load power demand exceeds the converter power capability, the FB pin voltage
exceeds VFBlin, the internal pull-up of the pin is disconnected and the pin starts sourcing
a 3 µA current that charges the capacitor C7 of the schematic in Figure 2.
As the voltage across the FB pin reaches the VFB_olp threshold (4.8 V typical), the VIPer25
stops switching (see Figure 26) and is not allowed to switch again until the VDD voltage has
fallen below VDD_RESTART (4.5 V typical) and risen again up to VDD_ON (14 V typical, see
Figure 25). Therefore the value of C7 defines the maximum duration of an overload event
without shutting down the IC.
If the overload (or short-circuit) is not removed, the protection is tripped again and the
system works in auto-restart mode (see Figure 25, 27, and 29).
If the overload disappears, the converter resumes normal working mode at the first VDD
recycling, as shown in Figure 29 and 30.
C7 is needed because usually the value of the C6 capacitor, coming from the loop stability
calculations, is too small to ensure an OLP delay time long enough to bypass the initial
output voltage transient at startup. The value of C7 can be chosen high enough to provide
the needed delay, and the value of R12 chosen so that the R12-C7 pole does not affect the
stability of the loop.
During an overload the converter is operated at a very low duty cycle, being the MOSFET
kept in the OFF state for most of the time. This results in a very low average power
throughput, which is safe for the power elements in this condition.
Figure 25. OLP: output short and protection
tripping
Figure 26. OLP: output short and protection
tripping - zoom
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Figure 27. OLP: steady-state (auto-restart
mode)
Figure 28. OLP: steady-state (auto-restart
mode) - zoom
Figure 29. OLP: short removal and restart
Figure 30. OLP: short removal and restart zoom
2.7
2nd OCP protection
The VIPer25 is provided with a first adjustable level of primary overcurrent limitation that
switches off the power MOSFET if this level is exceeded. This limitation acts cycle-by-cycle,
and its main purpose is to limit the maximum deliverable output power. A second level of
primary overcurrent protection is also present, which is not adjustable but fixed to 1.2 A
(typical value), to provide protection against short-circuit of the secondary rectifier, short circuit on the secondary winding, or a hard-saturation of the flyback transformer. If the drain
peak current exceeds this second overcurrent protection threshold, the device enters
a warning state. If, during the next ON time of the power MOSFET, the second level of
overcurrent protection is exceeded again, the device assumes that a secondary winding
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short-circuit or a hard-saturation of the transformer is occurring and stops the PWM activity.
To re-enable the operation, the VDD pin voltage must be recycled, which means that VDD
must go down to VDD(RESTART), then rise up to VDDon. At this point the MOSFET restarts
switching. If the cause of activation of the second overcurrent protection is still present, the
protection is tripped again and the system works in auto-restart mode, resuming normal
operation as soon as the cause of the fault is removed and the VDD recycled.
This protection has been tested in different operating conditions, short-circuiting the output
diode. The following figures show the behavior of the system during these tests.
With faults such as those described, the 2nd OCP protection of the VIPer25 stops the
operation after two switching cycles, therefore avoiding high currents in the primary as in the
secondary windings and through the power section of the VIPer25 itself. Figure 32 shows
the operation when a permanent short-circuit is applied on the secondary winding. Most of
the time the power section of VIPer25 is off, eliminating any risk of overheating.
Figure 31. 2nd OCP protection tripping
Figure 32. Operating with secondary winding
shorted - restart mode
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Figure 33. Operating with secondary winding
shorted - restart mode – zoom1
2.8
Figure 34. Operating with secondary winding
shorted - restart mode – zoom2
Voltage feed-forward function
As briefly explained in Section 2.1, in a quasi-resonant converter the switching frequency
increases with the input voltage.
According with Equation 3, for a fixed value of the maximum peak primary current, IPK_MAX,
the maximum deliverable output power is proportional to the switching frequency.
Equation 3
P IN
TRAFO MAX
1
= --- ⋅ L P ⋅ I
PK
2
MAX
2
⋅ f sw
If a power supply is designed in order to deliver a certain maximum output power at
minimum input voltage, its power capability at maximum input voltage can be twice or more
due to the increase of the switching frequency.
The system may not be able to detect an overload at high line because the load power
demand does not exceed the converter power capability, therefore leading to overheating of
the transformer, power MOSFET, and output diode.
In order to resolve this issue, the VIPer25 is provided with a voltage feed-forward function,
which acts by lowering the maximum drain current as the input voltage increases.
This is realized by connecting a resistor between the auxiliary winding of the transformer
and the ZCD pin (RFF resistor in Figure 35). In fact, during the ON time of the MOSFET, the
auxiliary winding voltage is negative and proportional to the flyback input voltage, therefore,
through the RFF resistor a current almost proportional to the input voltage is sunk by the
ZCD pin. Thanks to the adjustable current limitation VIPer25 function, the maximum peak
drain current is reduced proportionally to the increase of the input voltage value.
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Figure 35. Implementing input voltage feed-forward function
2&&2
#URRENTLIMITCOMPARATOR
$OVP$
2OVP2
!UXILIARY
WINDING
:#$PIN
2LIM
2
#URRLIM
",/#+
/60$%4%#4)/.
,/')#
4O07-LOGIC
&ROM3ENSE&%4
4O/60PROTECTION
!-
Figure 36 and Table 18 show the output current when the overload protection is tripped
(IOLP, blue line), while the IOUT_REST is the output current value at which the converter starts
working again once the overload protection has been tripped and the output load gradually
reduced.
Table 18.
IOUT_OLP and IOUT_REST values
VIN (VAC)
IOUT_OLP [A]
IOUT_REST [A]
90
2.12
1.88
115
2.26
1.97
230
2.37
1.96
265
2.31
1.88
Figure 36. IOUT_OLP and IOUT_REST vs. VIN_AC diagram
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2.9
AN3286
Output overvoltage protection
During the power MOSFET OFF time the voltage generated by the auxiliary winding tracks
the converter’s output voltage, through the transformer’s auxiliary-to-secondary turn ratio.
The diode D2 is forward biased, and the voltage divider made up of R14 and R3 (see
Figure 2 and Figure 35) between the auxiliary winding and the ZCD pin performs an output
voltage monitor function; if the voltage applied to the ZCD pin exceeds the internal VOVP
threshold (4.2 V typical) for four consecutive switching cycles, the controller recognizes an
overvoltage condition and shuts down the converter. This is done to provide high noise
immunity and avoid spikes erroneously tripping the protection. To re-enable operation the
VDD voltage must be recycled.
As the value of R3 has already been selected as a consequence of the maximum output
power that the converter must deliver, R14 can be chosen according to the following formula
to reach the desired output overvoltage threshold, VOUT_OVP:
Equation 4
R OVP
( R14 )
N AUX
1
= -------------------------------------------- ⋅ -------------- ⋅ ( V OU T
V OVP
N SEC
-------------------------–I
R LIM ( R 3) ZCD
OVP +
V ϒD4 – V ϒD2 ) – V OVP
where Vγ D2 and Vγ D4 are the forward drop of the diodes D2 and D4 respectively, NAUX and
NSEC are the auxiliary and secondary turns numbers respectively, and IZCD is the pull-up
current exiting the ZCD pin.
The above formula, solved for VOUT_OVP, gives about 6.5 V for the setting of the presented
board.
This value has been verified experimentally, short-circuiting the lower resistor (R9) of the
output voltage divider and therefore producing an output overvoltage, as shown in the
figures below. In Figure 37 it is possible to see that, as VOUT reaches the value of about
6.5 V, the converter stops switching. In the same figure the ZCD pin voltage and the FB pin
voltage are given. The crest value of the ZCD pin voltage during the MOSFET OFF time
tracks the output voltage. The converter is shut down as the ZCD pin voltage reaches the
VOVP threshold (4 V, within the tolerance), as shown in Figure 38.
Figure 37. Output overvoltage protection at
115 VAC 0.2 A
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Figure 38. Output overvoltage protection at
115 VAC 0.2 A - zoom
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The OVP protection is in auto-restart mode; after the shutdown for the protection tripping,
the VDD recycles and, if the overvoltage is still present, the protection is tripped again,
indefinitely, until the cause of the fault is removed, at which point the converter resumes
normal operation, as shown in Figure 39.
Figure 39. Auto-restart mode of the
overvoltage protection
2.10
Figure 40. Auto-restart mode of the
overvoltage protection - zoom
Brownout protection
Brownout protection is basically a not-latched device shutdown function activated when the
mains voltage falls below the minimum specification of normal operation. There are several
reasons why it may be desirable to shut down a converter during a brownout condition.
Firstly, to avoid overheating of the primary power section due to an excess of RMS current.
Secondly, to avoid spurious restarts during converter power-down, which may cause the
output voltage to not decay to zero monotonically. A typical example of this happens when
the mains is unplugged in a converter designed for the European range (230 VAC). The
converter is shut down due to the overload protection (due to the low input voltage the
converter is not able to supply the full power) but the voltage on the bulk capacitor is still
higher than VDRAIN RESTART, so the device starts again and the output voltage rises again.
This situation may be dangerous for some loads and in many applications it is advisable to
avoid it.
If this protection is required, the BR pin of VIPer25 must be connected to the DC HV bus
through a voltage divider. Otherwise the pin is to be connected to GND (the function is
disabled until VBR is lower than VDIS, which is 50 mV typ., as shown in Figure 41). In the
presented converter both settings are possible acting on the jumper J (see schematic in
Figure 2).
The BR pin is the inverting input of a comparator whose non-inverting input is internally
referenced to VBRth (0.45 V typ.). The PWM is disabled if the voltage applied at the BR pin is
below VBRth, and is enabled when the BR pin voltage is above VBRth + VBRhyst, where
VBRhyst is a voltage hysteresis (50 mV, typ. value) that ensures noise immunity.
The brownout comparator is also provided with current hysteresis; an internal current
generator, IBRhyst (9.5 µA typ.), is ON as long as the voltage applied at the brownout pin is
below VBRth and is OFF if the voltage is above VBRth + VBRhyst.
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Figure 41. Brownout function block diagram
6$)3
$)3!",%
"2
6"2HYST
6).?/+
6"2TH
)"2HYST
!-
The current hysteresis provides an additional degree of freedom, allowing the designer to
set the ON and OFF thresholds of the input voltage separately from each other, by properly
choosing the values of the resistors of the voltage divider on the BR pin. The following
relationships can be established:
Equation 5
VI N
OFF
R H + RL
= V BRth ⋅ ⎛ ---------------------⎞
⎝ RL ⎠
Equation 6
VI N
ON
RH + RL
= ( V BRth + V BRhyst ) ⋅ ⎛ ---------------------⎞ + R H ⋅ IBR hyst
⎝ RL ⎠
resulting in VIN_OFF = 83 V and VIN_ON = 114 V for the setting of the board. This can be
verified in the figures below, showing how brownout protection works in the VIPer25. For
a better understanding, DC input voltage has been applied.
At mains plug-in, the IBRhyst current generator is activated as soon as the drain voltage
exceeds VDRAIN_START, pulling down the BR pin voltage. If VIN < VIN_ON, VBR is lower than
0.45 V and the PWM is not activated, as shown in Figure 42. If VIN > VIN_ON, the BR pin
voltage is higher than 0.45 V, the PWM is enabled and the switching starts as the VDD
voltage reaches VDDon (Figure 43).
As the input voltage decreases, the IC continues switching until VIN > VIN_OFF, which means
VBR > 0.45 V (Figure 44). When VIN falls below VIN_OFF, the IBRhyst current generator is
turned on, therefore pulling the BR pin voltage below 0.45 V and shutting down the system.
The load is no longer supplied, and the output voltage decays monotonically to zero
(Figure 44).
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Figure 42. Input voltage steps from 60 VDC to
100 VDC - no turn-on
Figure 43. Input voltage steps from 60 VDC to
115 VDC - turn-on
Figure 44. Input voltage steps from 115 VDC to Figure 45. Input voltage steps from 115 VDC to
88 VDC - no turn-off
80 VDC - turn-off
2.11
EMI measurements
Pre-compliance tests to the EN55022 (Class B) European normative have been performed
using an EMC analyzer and an LISN.
The quasi-peak and average EMC measurements at 230 VAC full load have been performed
and the results are shown in Figure 46 and 47 respectively.
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Figure 46. Quasi-peak measurement at 230 VAC - full load
Figure 47. Average measurement at 230 VAC - full load
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2.12
Testing the board
Board layout
Figure 48. Complete layout
Figure 49. Top overlay
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Conclusions
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Figure 50. Bottom layer
3
Conclusions
The presented flyback converter is suitable for different applications. It can be used as an
external adapter or as an auxiliary power supply in consumer equipment. Special attention
was dedicated to light load performance and the bench results are good with very low input
power in light load conditions. The efficiency performances were compared with
requirements of the ENERGY STAR program (Version 2.0) for an external AC-DC adapter
with very good results; the measured active mode efficiency is always higher in respect to
the minimum required.
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Test equipment and measurement of efficiency and light load performance
Appendix A
Test equipment and measurement of
efficiency and light load performance
The converter input power has been measured using a wattmeter. The wattmeter
simultaneously measures the converter input current (using its internal ammeter) and
voltage (using its internal voltmeter). The wattmeter is a digital instrument, so it samples the
current and voltage and converts them into digital forms. The digital samples are then
multiplied, yielding the instantaneous measured power. The sampling frequency is in the
range of 20 kHz (or higher, depending on the instrument used). The display provides the
average measured power, averaging the instantaneous measured power in a short period of
time (1 s typ.).
Figure 51 shows how the wattmeter is connected to the UUT (unit under test) and to the AC
source, as well as the wattmeter’s internal block diagram.
An electronic load has been connected to the output of the power converter (UUT), enabling
the user to set and measure the converter load current, while the output voltage has been
measured by a voltmeter. The output power is the product of the load current vs. output
voltage.
The ratio between the output power, calculated as previously mentioned, and the input
power, measured by the wattmeter, is the converter’s efficiency. It has been measured in
different input/output conditions acting on the AC source and on the electronic load.
Notes on input power measurement
This section shows two possible connections between the wattmeter and the unit under test
(UUT) for power measurements, each represented in Figure 51 by the connection of the
switch either in position 1 or in position 2.
Figure 51. Connection of the UUT to the wattmeter for power measurements
If the switch shown in Figure 51 is in position 1 (see also the simplified scheme in
Figure 52), the ammeter’s internal shunt resistance (which is higher than zero) must be
taken into account. This resistance produces a voltage drop (then an input measured
voltage) higher than the input voltage at the UUT's input. This voltage drop is generally
negligible if the UUT's input current is low (for example, when measuring the input power of
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Test equipment and measurement of efficiency and light load performance
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the UUT at light load conditions), but at heavy load conditions, when the UUT input current
increases, the error introduced in the measurement with this setting can be relevant.
Figure 52. Suggested connection for low power measurements
(switch in position 1)
In this case, it is advisable to connect the switch shown in Figure 51 to position 2 (see
simplified scheme in Figure 52); the UUT’s input voltage is measured directly to the UUT’s
input terminal and the input current does not affect the measured input voltage.
Figure 53. Suggested connection for high power measurements
(switch in position 2)
With this setting, the measurement error is introduced by the shunt resistance of the
voltmeter, which is not infinite, that then causes a leakage current inside the voltmeter itself.
This current is measured by the ammeter together with the UUT’s input current, but the error
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Test equipment and measurement of efficiency and light load performance
is negligible at heavy loads, when the UUT’s input current is much higher than the
voltmeter's leakage current.
On the other hand, at light load conditions, when the UUT’s input current decreases and
approaches the voltmeter’s leakage current, the measurement error introduced with this
setting becomes significant.
To conclude, it is possible to say that the setting shown in Figure 52 should be used for light
loads and standby measurements, while the setting in Figure 53 for heavy loads and
efficiency measurements. If it is not clear which measurement scheme has the least effect
on the results, you can try with both and register the input power’s lower value.
As noted in IEC 62301, instantaneous measurements are appropriate when power readings
are stable. The UUT should be operated at 100% of the nameplate output current output for
at least 30 minutes (warm-up period) immediately prior to conducting efficiency
measurements. After this warm-up period, the AC input power should be monitored for
a period of five minutes to assess the stability of the UUT. If the power level does not drift by
more than 5% from the maximum value observed, the UUT can be considered stable and
the measurements can be recorded at the end of the five minute period.
If the AC input power is not stable over a five minute period, the average power or
accumulated energy should be measured over time for both the AC input and DC output.
Some wattmeter models allow integrating the measured input power in a time range and
then measuring the energy absorbed by the UUT during the integration time. The average
input power is calculated by dividing the measured energy by the integration time itself.
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References
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References
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1.
ENERGY STAR® Program Requirements for Single Voltage External Ac-Dc and Ac-Ac
Power Supplies (version 2.0)
2.
VIPer25 datasheet
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Revision history
Revision history
Table 19.
Document revision history
Date
Revision
26-Aug-2011
1
Changes
Initial release.
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